DISSERTATION. the Degree Doctor of Philosophy in the. Graduate School of The Ohio State University. Chieh Kai Yang, M.S.E.E.

Transcription

1 Trapping Effects in AlGaN/GaN HEMTs for High Frequency Applications : Modeling and Characterization Using Large Signal Network Analyzer and Deep Level Optical Spectroscopy DISSERTATION Presented in Partial Fulfillment of the Requirements for the Degree Doctor of Philosophy in the Graduate School of The Ohio State University By Chieh Kai Yang, M.S.E.E. Graduate Program in Electrical and Computer Engineering The Ohio State University 2011 Dissertation Committee: Prof. Patrick Roblin, Adviser Prof. Steve A. Ringel Prof. Siddharth Rajan

2 c Copyright by Chieh Kai Yang 2011

3 ABSTRACT Any defect site existing in the AlGaN/GaN HEMTs can be electrically active during device operation. The activated defect site not only could lead to a degradation in the output characteristics but may introduce additional nonlinearity which seriously downgrades the values of devices for various applications. This motivates us to study the detailed path experimentally and theoretically how an electrically-activated defect site could impact the device performances during practical device operation. In this study, the goal is (1) to give device engineers ideas on how further improvements can be devised to strengthen the existing GaN technology and (2) to provide circuit designers with better understanding on how to use GaN devices more efficiently for the development of reliable commercial GaN products for higher power applications in wireless systems. Single tone characterization results of AlGaN/GaN HEMTs for Class A operation are presented and compared. A new combined large signal network analyzer/deep level optical spectroscopy system is utilized to study the impact of illumination on the CW large-signal load line and small-signal S-parameters variations to identify the possible energy level of the trapping center responsible for the degradation of the device performance. A new pulsed-iv pulsed-rf coldfet technique is introduced to extract parasitic elements existing in the access regions of AlGaN/GaN HEMTs. The observation ii

4 of bias-dependence is detailed and a simple semi-physical model is proposed which provides a satisfactory description of experimental results. The low-frequency noise, an important figure of merit in terms of reliability, is briefly-reviewed. Additive phase noise measurements are presented and the effects of illumination and load impedance are examined. A physical expression is derived and simulated which successfully establishes a relationship between the access resistance and the low-frequency noise and provides a qualitative description of the measurement results. iii

5 This is dedicated to my family. iv

6 ACKNOWLEDGMENTS I would like to sincerely thank my adviser, Professor Patrick Roblin, for all of his time, effort, advice, and encouragement with my research and my entire studies. His insights and guidance have made many research works to be done successfully and finally this dissertation remarkably better. Above all, he has provided me with an excellent role model as a researcher, a teacher, and a scholar, so that I have tremendous respect for him. Together with Professor Roblin, Professor Steve A. Ringel, Professor Siddharth Rajan, and Professor Waleed Khalilcguided me through the dissertation phase of my studies by forming a committee. I would like to express thanks to them for the dedication to many hours of their time to review my work. They always enriched my work with their valuable feedback and suggestions. It is also a pleasure to thank visiting scholar: Dr. Fabien De Groote. He not only motivated me to enjoy experimental research works, but also gave me many productive discussions and guidance. I would like to express my great thankfulness for colleagues in Professor Steve A. Ringel s group. In particular, Dr. Aaron Arehart and Andrew Malonis, for the great help of setting up the deep level optical spectroscopy system and providing valuable advices when experimental problems are encountered. They were of great assistance to me in the lab operations. v

7 I also would like to express my gratitude for my colleagues in our Non-Linear RF Lab. In particular, Dr. SeokJoo Doo, for the great help of learning and utilizing the LSNA and many other experimental systems. Inwon Suh, Young-seo Ko, and Haedong Jang we shared good times, and thanks for your kind help in the Lab. They were of great assistance to me in this research. I also would like to thank Office of Naval Research for the support to this study. I have taken a great chance of studying at OSU and completed the work successfully with their kind understanding and support. I also would like to take the opportunity to thank my Mother who sacrificed so much for me to be here. The support and guidance through my entire life has shaped me into the individual I am today. vi

8 VITA June 16, Born - Miaoli, Taiwan May 22, B. S. Electrical Enigneering, National Tsin Hui University, Hsinchu, Taiwan May 10, M. S. Electrical Enigneering, National Tsin Hui University, Hsinchu, Taiwan September present Ph. D. Student, Electrical and Computer Engineering, The Ohio State University. January March Graduate Research Associate, Electrical and Computer Engineering, The Ohio State University. PUBLICATIONS Research Publications C.-K. Yang, P. Roblin, F. De Groote, S. Ringel, S. Rajan, J. P. Teyssier, C. Poblenz, Y. Pei, J. Speck and U. Mishra, Pulsed-IV Pulsed-RF Cold-FET Parasitic Extraction of Biased AlGaN/GaN HEMTs using Large Signal Network Analyzer. IEEE Transactions on Microwave Theory and Techniques, vol.58, p.1077 V 1088, May Inwon Suh, Patrick Roblin, Youngseo Ko, Chieh-Kai Yang, Andrew Malonis, Aaron Arehart, Steven Ringel, Christiane Poblenz, Yi Pei, James Speck, and Umesh Mishra, Additive Phase Noise Measurements of AlGaN/GaN HEMTs Using a Large Signal Network Analyzer and a Tunable Monochromatic Light Source. ARFTG 74th Conf., Nov vii

9 C.-K. Yang, P. Roblin, A. Malonis, A. Arehart, S. Ringel, C. Poblenz, Y. Pei, J. Speck and U. Mishra, Characterization of Traps in AlGaN/GaN HEMTs with a Combined Large Signal Network Analyzer/Deep Level Optical Spectrometer System. IEEE MTT-S Int. Symp., vol. 10, no. 3, May F. De Groote, P. Roblin, J. P. Teyssier, C. Yang, S. Doo, M. Vanden Bossche, Pulsed Multi-Tone Measurements for Time Domain Load Pull Characterizations of Power Transistors. ARFTG 73th Conf., vol. 16, no. 12, pp , Dec Chieh-Kai Yang, Chia-Ming Yang, Hua-Hsien Liao, and Sheng-Fu Horng, Hsin-Fei Meng, Current injection and transport in polyfluorene. Appl. Phys. Lett., no. 91, , FIELDS OF STUDY Major Field: Electrical and Computer Engineering Studies in Microwave and Microelectronics: Prof. Patrick Roblin viii

10 TABLE OF CONTENTS Page Abstract Dedication Acknowledgments Vita List of Tables ii iv v vii xi List of Figures xii Chapters: 1. Introduction Characteristics of GaN Technology and Trap-related Problems Degradations in Device Performances Current Collapse Frequency Dispersion Gate Lag and Drain Lag Low-frequency Noise Traps Distributions and Technology Improvements Motivations and Contributions of this Thesis Dissertation outline Large- and Small-signal Characterizations of GaN HEMTs with Optical Spectroscopy Introduction Fundamental Power Amplifier Characterization Parameters ix

11 2.3 Single-tone Characterizations of GaN HEMTs for Class-A Operations Characterizations with Optical Spectroscopy CW-RF Loadlines Measurement Time evolution CW-RF loadline measurements S-parameters measurement Summary Quasi Static Device Parasitics Introduction Extraction of Parasitic Components Bias-dependent Parasitic Components Parasitic Capacitance Parasitic Resistance Modeling of the Parasitic Resistance Parasitic Inductance Modeling of the Parasitic Inductance Summary Low-frequency Noise in GaN HEMTs Introduction Shockley-Read-Hall Statistics and Random Telegraph Signal Shockley-Read-Hall Statistics Random Telegraph Signal Low-frequency Noise and Cyclostationary Effect Low-frequency Noise Cyclo-stationary Effect Additive Phase Noise Measurements Access Resistance and Low-frequency Noise in AlGaN/GaN HEMTs Access Resistance and Low-frequency Noise MATLAB Simulations Summary Conclusions and Future Works Conclusions Future Works Appendix A. Pulsed-IV Pulsed-RF Measurement Setup Bibliography x

12 LIST OF TABLES Table Page db gain compression point P in,ava, 1dB versus drain bias voltage for both unpassivated and passivated GaN HEMTs Time constants of the variation of current swing for different energies of illuminations Extracted average values of C pgs, C pds, and C pgd are used for both unpassivated and passivated devices Reference parasitic resistances at the source and drain sides measured at the quiescent bias of V GS = 0 V and V DS = 0 V Reference parasitic inductances at the source and drain sides measured at the quiescent bias of V GS = 0 V and V DS = 0 V xi

13 LIST OF FIGURES Figure Page 1.1 Characteristics of the GaN semiconductor compared with other semiconductor materials such as Si, GaAs, and SiC. [ Source : Nikkei Electronics Inc. ] Comparisons of material properties for different semiconductor technologies [2] Possible distributions of traps residing in the conventional GaN HEMT structure Device structures of AlGaN/GaN HEMTs with and without SiN passivation layer over the AlGaN surface prepared by UCSB group Schematics of the polarization directions and band diagram of a Gaface AlGaN/GaN HEMT Schematic of a PA operation. P in,ava, P DC, P out, and P diss are the available input power, available DC supply power, output power delivered to an external load, and power dissipation, respectively A typical PA performance exemplifying the nonlinear behavior as available input power continues to grow in terms of the output power P L,out, the power gain G L, and the power-added efficiency P AE Schematic of the output power P L,out of a PA subject to single tone excitation showing fundamental frequency (solid line), second (dash line), and third (dash-dot line) harmonic components Schematic of the measurement system in the single-tone characterization of GaN HEMTs using a Large-signal Network Analyzer(LSNA).. 18 xii

14 2.5 Measured output loadlines at (V GS, V DS )=(-3 V, 10V) (red), (-3 V, 15V) (blue), and (-3 V, 20V) (green) with corresponding load impedances equal to 50Ω, 133Ω, and 220Ω, respectively, in comparison with DCIV curves (dashed line) Measured output power P L,out versus available input power P in,ava at fundamental frequency (red), second (blue), and third (green) harmonic components. Four quiescent bias points are examined at (V GS, V DS )=(-3 V, 10V) (solid line), (-3 V, 15V) (dotted line), and (-3 V, 20V) (dashed line) Measured power gain G L and power added efficiency P AE versus available input power P in,ava at fundamental frequency. Four quiescent bias points are examined at (V GS, V DS )=(-3 V, 10V) (solid line), (-3 V, 15V) (dotted line), and (-3 V, 20V) (dashed line) Measured average drain current I DS,avg versus available input power P in,ava. Four quiescent bias points are examined at (V GS, V DS )=(-3 V, 10V) (solid line), (-3 V, 15V) (dotted line), and (-3 V, 20V) (dashed line) Measured power dissipation P diss versus available input power P in,ava. Four quiescent bias points are examined at (V GS, V DS )=(-3 V, 10V) (solid line), (-3 V, 15V) (dotted line), and (-3 V, 20V) (dashed line) Experimental setup of LSNA/DLOS combined system Comparison of measured pulsed-ivs and RF loadlines at 2 GHz with and without illumination Photoionization spectrum of the variation of current swing for SiNpassivated and unpassivated AlGaN/GaN HEMTs Photoionization spectrum of average drain current for SiN-passivated and unpassivated AlGaN/GaN HEMTs Time evolution of the variation of current swing subject to 3.54 ev and 3.10 ev illuminations for unpassivated AlGaN/GaN HEMT xiii

15 2.15 Variation of S-parameters subject to different energy of illuminations with frequency sweeping from 1 GHz to 13 GHz for unpassivated Al- GaN/GaN HEMTs Variation of S-parameters subject to different energy of illuminations with frequency sweeping from 1 GHz to 13 GHz for unpassivated Al- GaN/GaN HEMTs Photoionization spectrum of the variation of S 21 magnitude at 2 GHz for SiN-passivated and unpassivated AlGaN/GaN HEMTs Photoionization spectrum of the variation of S 22 magnitude at 2 GHz for SiN-passivated and unpassivated AlGaN/GaN HEMTs Variation of the extrinsic transconductance g M and extrinsic drain conductance g D with different photon energy extracted from real part of Y 21 and Y 22, respectively, for SiN-passivated and unpassivated Al- GaN/GaN HEMTs Wave-equation based equivalent circuit used for the cold FET parasitics extraction Equivalent circuit under pinched-off condition used for the extraction of parasitic capacitance Example of measured (red dot) and fitted (black line) S-parameters for one DC bias point for the passivated device where S 12 and S 21 have been magnified five times larger Measured C pgs,t of unpassivated (circle mark) and passivated (square mark) devices for different DC bias points Measured C pds,t of unpassivated (circle mark) and passivated (square mark) devices for different DC bias points Measured C pgd,t of unpassivated (circle mark) and passivated (square mark) devices for different DC bias points xiv

16 3.7 Deviation of drain resistance R D = R D R D0 : measured (circle line for the unpassivated device and square line for the passivated device) and fitting (dash-dot line for the unpassivated device and dotted line for the passivated device) deviations of parasitic resistances on the drain side Deviation of source resistance R S = R S R S0 : measured (circle line for the unpassivated device and square line for the passivated device) and fitting (dash-dot line for the unpassivated device and dotted line for the passivated device) deviations of parasitic resistances on the source side Deviation of resistance: measured (circle line for the unpassivated device and square line for the passivated device) R S/L at drain and source sides with T = T T 0 where T 0 = 300K. Dashed line is the fitting result using Eq. 3.9 to determine α and R th in the equation Calculated change of the channel temperature with different quiescent bias points (circle line for the unpassivated device and square line for the passivated device) DC-IV curve and channel modulation index β ( calculated using coefficients for the drain side ) contour for unpassivated device DC-IV curve and channel modulation index β ( calculated using coefficients for the drain side ) contour for passivated device Example of measured (red dot) and fitted (black line) S-parameters for one DC bias point for the unpassivated device for V GS = 6 V Deviation of drain inductance L D = L D L D0 measured for the unpassivated (red circle line) and passivated devices (blue square line) and fitting results for the unpassivated device (red dash-dot line) Deviation of source inductance L S = L S L S0 measured for the unpassivated (red circle line) and passivated devices (blue square line) and fitting results for the unpassivated device (red dash-dot line) Equivalent circuit for modeling the effective reduction in parasitic inductances L G and L S observed xv

17 4.1 The electron and hole capture and emission processes involved in Shockley- Read-Hall Statistics considering one trapping center where E c being the conduction band edge, E v being the valence band edge, E i being the intrinsic Fermi-level, and E T being the energy level of the trapping center Schematic representation of the channel current RTS (I ds ) in a n- MOSFET. The time spent in the high current level is denoted as τ 1 whereas the time spent in the low current level is denoted as τ 0. The amplitude of the variation of the channel current with time is denoted as δi ds Possible energy and spacial locations of trapping centers in a AlGaN/GaN HEMT in equilibrium leading to 1/f low-frequency noise Comparison of additive phase noise on various illumination energies between different DC biasing conditions V DS = 10 V (solid line) and V DS = 15 V (dashed line) Comparison of phase noise dependence between passivated and unpassivated device on load impedances Qualitative illustration of variations of S δid in terms of the occupancy function of surface states f T due to the variation of photon energy Qualitative illustrations of variations of S δid in terms of the occupancy function of surface states f T due to the variation of load impedance R L for both unpassivated and passivated devices Pulsed-IV/RF measurement system utilizing the modified LSNA Measured traveling waves of the pulsed-rf signal with 700 ns and 2000 ns pulse widths are compared with those obtained for a CW-RF signal of the same peak power xvi

18 CHAPTER 1 INTRODUCTION 1.1 Characteristics of GaN Technology and Trap-related Problems In the early days of wireless development[1] vacuum tube devices have dominated for high power amplification at radio frequencies (RF). Continuing research activities in this field driven by the need for mobile and personal communications systems, have led to a variety of RF solid state devices with diverse device structures and materials, including Heterojunction Bipolar Transistors (HBTs), Metal Oxide Semiconductor Field Effect Transistors (MOSFETs), Laterally Diffused Metal Oxide Semiconductor FETs (LDMOSFET), Metal Semiconductor FETs (MESFETs) and High Electron Mobility Transistors (HEMTs). Most of these devices are constructed using the wellestablished Si and GaAs technologies. The recent trend towards higher and higher power densities for RF devices is fuelling further growth in the semiconductor industry. While the limits of the conventional technologies have begun to emerge, further demands for improved device performance is continuing increase and new device concepts and technologies are 1

19 emerging which promise to fulfill the enhanced requirements for the wireless applications. Within the many potential technologies currently available, GaN semiconductors, traditionally used in LEDs and laser diodes, are gradually appearing in a host of new applications of power devices. The key features offered by GaN semiconductors are listed in Fig. 6.1 along with comparisons with other semiconductor materials. Figure 1.1: Characteristics of the GaN semiconductor compared with other semiconductor materials such as Si, GaAs, and SiC. [ Source : Nikkei Electronics Inc. ] Compared to other semiconductors such as Si, GaAs, and SiC, GaN offers some critical characteristics appropriate for power device applications in the wireless system including higher breakdown strength, higher operating temperature, higher maximum current density, and etc. These advantages inherently come from the superior material properties of GaN semiconductors as illustrated in Fig

20 Figure 1.2: Comparisons of material properties for different semiconductor technologies [2]. In recent years, excellent performances have been demonstrated in the GaN-based devices. However, the issues of the nonlinearity and the reliability remain the major problem hampering the commercialization of those devices. Although the growth of high-quality nitride materials continue to advance by big steps, the nitride epitaxial layer is generally grown on SiC or sapphire substrate due to the lack of large area GaN substrates. Such substantial mismatch in the lattice constant and thermal conductivity between the substrate and the epitaxial layer has resulted in a significant biaxial strain leading to many extended defects. Moreover, impurities or lattice defects can be incorporated during the epitaxial growth process. The defect site introduced by any physical process mentioned above can be electrically active during device operation which not only could cause a degradation in the device characteristics but may also exacerbate the nonlinearity limiting the practical use of GaN-based devices. Some commonly-observed degradations in GaN-based devices are introduced in the following section. 3

21 1.2 Degradations in Device Performances Current Collapse Current collapse, an observation of the reduction of the dc drain current and the distortion of the dc current-voltage (I V ) characteristic, is usually viewed as a hot-carrier effect that is mostly observed in small device structures where a high applied electric field is common. The first current collapse in the AlGaN/GaN HEMT was first observed in [3]. The effect of current collapse is to increase the drain knee voltage (the minimum allowable drain voltage for linear operation) and to decrease the maximum achievable saturation drain current. The overall effect is that of a degradation of the available output RF power Frequency Dispersion The reduction in device output power due to current collapse is enhanced during high frequency operation and is further exacerbated when a large input signal is applied at the gate in order to increase power added efficiency (PAE) of the power transistor. This effect, named as frequency dispersion, is another mechanism that limits the device output power[7]. The measured output power of the device would be smaller than that predicted by the dc I V characteristic. Such a mechanism is also one of the main reasons that causes the discrepancy between the measured dc I V and pulsed I V characteristics since pulsed I V characteristic is usually measured with large pulse amplitudes and pulse width smaller than µs range. 4

22 1.2.3 Gate Lag and Drain Lag In addition to the power degradation problem, trapping effects also play a dominant role in the device transient response especially for applications demanding fast switching. The two major transient response problems observed in GaN HEMTs include gate and drain lagging effects. Gate lag is a delayed response of the drain current with respect to the variation of the applied gate voltage[8] and is usually related to the trapping mechanism in the vicinity of the gate contact. The dynamics of trapping mechanism is determined by the capture and emission processes of the trapping center within the bandgap. On the other hand, drain lag is a delayed response of the drain current with respect to the variation of the applied drain voltage. Drain lag is also a trapping-related mechanism which usually occurs in the vicinity of conducting channel. It can also be induced through the trapping mechanism resulting in gate lag[9]. A broad range of characteristic times for the dynamics of gate and drain lags have been observed which varies from nanoseconds to hundreds of seconds[8][9] Low-frequency Noise Low-frequency noise, also called flicker noise, which manifests itself at low frequencies (usually between 0.01 khz to 100 khz) with 1/f γ spectral density, is an important figure-of-merit for semiconductor devices which imposes a lower limit on the signal level and determines the sensitivity of a communication system. Studies of low-frequency noise in electron devices have shown that the noise source often arises from material defects and different types of localized states, which act as carrier trapping and detrapping centers[10][11][12]. The noise level in a semiconductor material or device is usually characterized by the dimensionless Hooge parameter[70]. Early 5

23 Surface states at the dielectric/algan Barrier interface Source Gate Drain Interface states at the Gate Contact/AlGaN Buffer layer interface Interface states at the AlGaN Buffer/Nucleation layer interface AlGaN Barrier 2DEG Channel GaN Buffer AIN Nucleation Layer Substrate Bulk traps inside the AlGaN Barrier Interface states at the AlGaN Barrier/Channel interface Bulk traps inside the GaN Buffer Figure 1.3: Possible distributions of traps residing in the conventional GaN HEMT structure. investigations of low-frequency noise in gateless AlGaN/GaN HEMTs have shown a high noise level with Hooge parameter on the order of With improvements in the material growth technology, a number of lower Hooge parameters, on the order of 10 4 and 10 3, in different GaN-based HEMT structures have been demonstrated[12]. However, reliability, usually indicated by an increase of low frequency noise, remains a major problem even with technology improvements which hampers the practical usage of GaN HEMTs in power transistor applications. 1.3 Traps Distributions and Technology Improvements Different kinds of trapping centers can reside in various locations of the GaN devices, as illustrated in Fig They are either introduced during the device growth process or electrically activated during normal device operation. Device improvements 6

24 have been the object of intensive research in order to suppress performance degradations cause by potentially-existing trapping centers. Technology improvements using silicon nitride passivation[14], capped-gan layer[15], or other materials[16] have shown the successful suppression of device degradations in AlGaN/GaN HEMTs associated with surface states. The concept of field plate is also utilized to reduce the high electric field peaking at the gate edge [17][16] which is ascribed to be the cause of the hot-electron-induced trapping effect[14][18]. The insertion of AlN layer between AlGaN and GaN layers[19] has demonstrated an improved channel mobility due to a decrease in alloy disorder scattering caused by interface states. Device structures with an inserted dielectric, insulator, or oxide layer between the gate metal and the AlGaN epitaxial layer has been proposed for AlGaN/GaN HEMTs[20] and demonstrated to exhibit better performances and reduced gate leakage current by reducing the impact of surface states or interface traps. Moreover, device structures using a thick AlN(>50nm) nucleation layer sandwiched between GaN buffer and SiC substrate along with the intentional carbon doping have been shown to yield reduced leakage current in the GaN buffer layer[21]. 1.4 Motivations and Contributions of this Thesis Excellent performances have been demonstrated by optimized AlGaN/GaN HFETs reported in the previous section. However, the degradations of nonlinearity and reliability over time remain the major issues even for devices with such technology improvements. This motivates us to study both experimentally and theoretically the physical process by which a electrically-activated defect site can impact the device performances 7

25 during practical device operation. The goal in this study is (1) to give device engineers ideas on how further improvements can be devised to strengthen the existing GaN technology and (2) to provide circuit designers with better understanding on how to use GaN devices more efficiently for the development of reliable commercial GaN products for higher power applications in wireless systems. Contributions of this thesis include 1) the development of a new combined large signal network analyzer/ deep level optical spectroscopy system for characterizing potential trap activities during practical device operation, 2) the utilization of a new pulsed-iv pulsed-rf technique to extract parasitic elements residing in the access regions of GaN devices. A semi-physical model is proposed which gives a satisfactory description of the experimental observation, 3) and the use of a new combined additive phase noise measurement system/ deep level optical spectroscopy to characterize the low-frequency noise in GaN devices. A physical model is derived which successfully explains the variation of the low-frequency noise in terms of the illumination and the load impedance variations. The devices under test in this dissertation were prepared by the UCSB group as shown in Fig. 1.4 which are grown at 720 C by PAMBE with an ohmic contact (Ti/Al/Ni/Au) and a Gate metal (Ni/Au/Ni). The gate length is 0.7 µm and the gate-source and gate-drain spaces are 0.5 µm and 1 µm, respectively. A schematic of the band diagram of the AlGaN/GaN HEMT is illustrated in Fig. 1.5 which shows the polarized charges, two-dimensional electron gas, charge-balanced surface states and interface charges[16]. 8

26 Figure 1.4: Device structures of AlGaN/GaN HEMTs with and without SiN passivation layer over the AlGaN surface prepared by UCSB group. Figure 1.5: Schematics of the polarization directions and band diagram of a Ga-face AlGaN/GaN HEMT. 9

27 1.5 Dissertation outline The dissertation is arranged as follows: 1) Characteristics of GaN semiconductors and encountered degradations are briefly introduced in this chapter. 2) Chapter two presents single tone characterization results for Class A operation. A new combined large signal network analyzer/ deep level optical spectroscopy system is utilized to study the impact of the illumination on the large-signal load line variation as well as small-signal S-parameters to identify the possible energy level of the trapping center which degrades the device performance. 3) In chapter three, a new pulsed-iv pulsed-rf technique is used to extract the parasitic elements locating in the access regions of AlGaN/GaN HEMTs. A simple semi-physical model is proposed which gives a satisfactory description of the measurement result. 4) Conventional lowfrequency noise theory is briefly reviewed in the beginning of chapter four. A new combined additive phase noise measurement system/ deep level optical spectroscopy is developed and characterization results of low-frequency noise in AlGaN/GaN HEMTs are presented. A physical expression is derived and simulated which establishes a close link between the low-frequency noise and the variation of the access resistance. 5) Finally, the results and conclusions obtained in this thesis are summarized and future continuation investigations are proposed. 10

28 CHAPTER 2 LARGE- AND SMALL-SIGNAL CHARACTERIZATIONS OF GAN HEMTS WITH OPTICAL SPECTROSCOPY 2.1 Introduction Among various applications of GaN devices, one of the most important utilizations is its use as a power amplifier(pa) for wireless communications. The major task of a PA is to amplify the incoming signal at the input port to a specified signal level at the output port for further transmission. In contrast to a conventional operational amplifier(opamp), which can be described as a linear system and has well-defined small-signal parameters for performance characterizations, a PA is usually operated in the large-signal regime not only to enhance its output power capabilities but also to increase the power-added efficiency(pae). A PA should therefore to be considered as a nonlinear system component, whose large-signal operating conditions often lead to detrimental effects on the output signal, resulting at the output in a distorted amplified replica of the input. On the other hand, it is well known that by depositing a SiN-passivation layer over the AlGaN surface of GaN HEMTs, degradations in small-signal parameters, such as 11

29 transconductance g m and cut-off frequency f T can be largely reduced[14]. Experimentally, photoionization can be used to recover the degraded device performances through a photon excitation process which has been demonstrated to effectively reduce the current collapse and the frequency dispersion in GaN HEMTs [3]. Therefore, by characterizing large-signal behaviors and small-signal parameters in combination with the photoionization method at the same time, we may be able to identify the possible physical sources causing device degradations in GaN HEMTs. In the first section, the fundamental characterization parameters for a power amplifier are introduced. Single-tone characterizations of GaN HEMTs with and without SiN-passivation layer are presented and compared in section two. Section three gives the optical spectroscopy results in terms of large-signal loadline variations. Finally, S-parameters are measured under illumination and resulting device characteristics analyzed. 2.2 Fundamental Power Amplifier Characterization Parameters Simply put, a PA is a power conversion device which converts the available DC supply power P DC to the high-frequency version of output power. This illustrated in Fig. 6.1 which present the fundamental concept of the PA operation. During PA operation, the available input power, P in,ava, is the input power reaching the input port of a PA from an external signal source and is defined by P in,ava (f) = 1 2 R{V in I in} (2.1) for a specified frequency f. The output power from a PA, P L,out, on the other hand, is the power exiting its output port and being delivered to an external load for the 12

30 P DC P in,ava PA P L,out P diss Figure 2.1: Schematic of a PA operation. P in,ava, P DC, P out, and P diss are the available input power, available DC supply power, output power delivered to an external load, and power dissipation, respectively. same specified frequency as the input signal defined by P L,out (f) = 1 2 R{V out I out} (2.2) The power gain, G L, at the specified frequency is the ratio between P L,out and P in,ava G L (f) = P L,out P in,ava (2.3) Due to the broad dynamic range of signals practically encountered in a PA, power quantities are usually expressed in logarithmic scale. When a reference to 1 mw is used, the unit of the power level is specified in dbm as: ( ) P P dbm =10 log 1mW (2.4) and the power gain in logarithmic scale is expressed as G L,dB =10 log (G L ) = P L,out,dBm P in,ava,dbm (2.5) Ideally, all the available DC supply power should be converted to the high frequency output power without any power loss. However, in practice, power loss or 13

31 power dissipation always occurs and this can be characterized by the power-added efficiency η add, η add = P L,out P in,ava P DC (2.6) in which P DC is calculated as the multiplication of the average DC bias drain voltage V DS and the average DC bias drain current I D P DC = V DS I D = V DS,avg I D,avg (2.7) where... denotes the time average value of the specified quantity. From the point of view of energy conservation, a power balance equation describing a PA operation can be written as P in,ava + P DC =P L,out + P diss (2.8) The power dissipation component P diss generated during PA operation is considered as a power conversion loss which usually dissipates itself in the form of an additional heat generation inside the PA and can be calculated by Eq. 2.8 after obtaining all the information needed for the various components featured in this expression. Based on Eq. 2.6 and Eq. 2.8, the ultimate goal of a power amplifier is to maximize the power-added efficiency while minimizing the power dissipation at the same time for a given DC supply power. This the major challenge faced by device and circuit design engineers. Now, by combining Eq. 2.3 and Eq. 2.6, we can derive that η add = P in,ava (G L 1) P DC (2.9) Eq. 2.9 implies that, for a given DC supply power, increasing the input power while maintaining the power gain much greater than one gives a feasible way to enhance 14

32 the P AE performance of a PA. As a result, it is reasonable to let the PA operate in the large-signal regime in order to get better P AE performance. However, due to the inherent limitation of practical power amplifiers (either due to the voltage limitation or the current limitation), strong nonlinearity shows up as the input power entering the PA continues to grow. Fig. 6.2 illustrates such nonlinear behavior arising during PA operation when the available input power is gradually increased. P L,out,dBm P L,out, 1dB,dBm 1 db G L,dB G L, 1dB add P in,ava, 1dB,dBm P in,ava,dbm Figure 2.2: A typical PA performance exemplifying the nonlinear behavior as available input power continues to grow in terms of the output power P L,out, the power gain G L, and the power-added efficiency P AE. Assuming for simplicity that the power amplifier can be described using a memoryless nonlinear system expression with up to third-order approximation, the signal y(t) at the output of the nonlinear amplifier is given by: y (t) =k 1 x (t) + k 2 x 2 (t) + k 3 x 3 (t) (2.10) with x(t) being the input signal applied to the PA and k i with i = 1, 2, 3 being the coefficients corresponding to first-, second-, and third-order response, respectively. 15

33 The power gain G L dependence of P in,ava can be derived by[2] G L (P in,ava ) =G L (P in,ava 0) ( ) 2 2 k3 P in,ava (2.11) k 1 Usually k 3 /k 1 is negative, therefore, the above derivation accounts for the large-signal gain compression, i.e. a decrease of the ideal constant value G L (P in,ava 0), which can be characterized by the 1 db gain compression point as indicated in Fig. 6.2 by the red dot. From a physical point of view, the gain compression behavior under large-signal excitation induces an another kind of power conversion loss. Indeed rather than dissipating itself under the form of additional heat, part of the DC power is transformed to higher-order RF harmonics due to the nonlinear characteristics of the power amplifier. For a third-order nonlinear system, Fig. 4.3 illustrates how the converted RF power can grow at the n-th harmonic components at a growth rate of n db per db increase in the input power. The corresponding output powers generated at harmonics P L,out,2f and P L,out,3f in this case are then given by[2] P L,out,2f = 1 2 G L P L,out,3f = 1 4 G L k 2 k 1 k 3 k 1 2 P in,ava (2.12) 3 P in,ava (2.13) Practically, multi-tones or a band-limited signal are more often encountered during practical PA operation. More complex PA performance characterizations can be referenced in [2] while only single tone excitation will be considered in this dissertation. 16

34 P L,out,dBm P L,out, 1dB,dBm 1 db P L,out,1f P L,out,2f P L,out,3f P in,ava, 1dB,dBm P in,ava,dbm Figure 2.3: Schematic of the output power P L,out of a PA subject to single tone excitation showing fundamental frequency (solid line), second (dash line), and third (dash-dot line) harmonic components. 2.3 Single-tone Characterizations of GaN HEMTs for Class- A Operations Single-tone characterization is easy to be implemented and can provide an immediate indication of device performances, such as the output power, the power gain, the linearity, and the PAE. For the sake of simplicity, Class-A operation is selected for the power amplifier since other types of PA operations usually require harmonic tuning techniques[2]. Fig. 4.4 shows the experimental setup in the single-tone characterization of GaN HEMTs with fundamental frequency of the signal source equal to 2 GHz where the tuner at port 1 is tuned to the conjugate matching condition enabling the maximum power transfer from the signal source to the DUT (Device Under Test). The input power P in,ava is equal to 16 dbm which stands for a gate-source voltage V gs variation from -6 V(pinched-off) to 0 V. 17

35 Large Signal Network Analyser Signal Source Port1 Tuner DUT Tuner Port2 Current Sensor Current Sensor Bias Tee Bias Supply Bias Tee Figure 2.4: Schematic of the measurement system in the single-tone characterization of GaN HEMTs using a Large-signal Network Analyzer(LSNA). The measured large-signal loadlines of GaN HEMTs with and without SiN-passivation along with corresponding DCIV curves are presented in Fig Three DC bias points are examined with gate bias voltage V GS equal to -3 V and drain bias voltage V DS increased from 10 V to 20 V. The load impedances are chosen such that the extensions of the loadlines at different DC bias points are reached at the same knee voltage and maximum drain current. For the device with SiN-passivation as shown in Fig. 4.5, in comparison with the measured DCIV curves, the loadline at lower drain bias voltages (10 V) fails to reach the maximum drain current at the knee voltage as predicted by the measured DCIV. Such deviation indicates a drain current dispersion due to the frequency dispersion phenomenon mentioned in chapter one when a large RF signal is applied to the device. In addition, at higher drain bias voltage with larger load impedance, the dispersion effect is found to be exacerbated which may be explained by the so-called IV knee walk-out reported in [4] where the degradation was mainly ascribed to thermal effects. Such similarity suggests that the thermal effect may be one of the physical sources responsible for the frequency dispersion observed in the passivated GaN HEMT. 18

36 When the same characterization conditions are applied to the unpassivated device, a significant dispersion effect is observed. A large deviation between the measured loadline and DCIV curves is identified even at lower drain bias voltage (10 V) as shown in Fig In comparison with the passivated device, our observations indicate that a source of dispersion other than thermal effects may be superimposed upon the unpassivated device and causes the additional current dispersion. Such additional dispersion may be due to the trapping effects commonly encountered in many Nitridebased devices[3] DCIV 0.1 Unpassivated V GS = 0 V 0.12 DCIV 0.1 Passivated V GS = 0 V I DS (A) ( 3V,10V) ( 3V,15V) ( 3V,20V) I DS (A) ( 3V,10V) ( 3V,15V) ( 3V,20V) Ω 133 Ω 220 Ω 50 Ω 133 Ω 220 Ω V (V) DS V (V) DS Figure 2.5: Measured output loadlines at (V GS, V DS )=(-3 V, 10V) (red), (-3 V, 15V) (blue), and (-3 V, 20V) (green) with corresponding load impedances equal to 50Ω, 133Ω, and 220Ω, respectively, in comparison with DCIV curves (dashed line). As mentioned in the previous section, GaN devices in the application of the power amplifier is better described by a nonlinear system. Therefore, it is desirable to characterize its nonlinearity during large-signal operation. The measured output power P L,out versus available input power P in,ava at fundamental frequency and harmonic components for both unpassivated and passivated GaN HEMTs are presented 19

37 in Fig. 4.6 in which P L,out,fun represents the output power delivered to an external load at fundamental frequency. As expected, higher P L,out,fun is observed in the passivated device possibly due to smaller dispersion effect compared to the unpassivated counterpart. In addition, owing to a larger available drain voltage swing allowable at larger drain bias voltage, P L,out,fun increases at the same time even with a little reduction in the maximum current swing due to dispersion effect until these two effects are comparable to each other. Also demonstrated in Fig. 4.6 is an increase of P L,out,fun with gradually increasing input power. However, the increasing input power also causes higher power distribution over the various harmonic components. Measured output powers of P L,out,2nd and P L,out,3rd indicate a worsened dispersion effect happening in the unpassivated device and simultaneous larger nonlinearities. Moreover, it can be noted that while the passivated device gives a negligible variation of P L,out,2nd and P L,out,3rd in terms of a change with the drain bias voltage, a noticeable drain-bias-dependence of the output power of the harmonics can be identified in the unpassivated device. Such observation indicates an additional nonlinearity is introduced at higher drain bias voltage. Besides measuring the output power at harmonic components, the 1 db gain compression point P in,ava, 1dB is another important figure of merit in the nonlinearity characterization. Fig. 4.7 gives the calculated power gain G L and power added efficiency PAE for both unpassivated and passivated GaN HEMTs in terms of the variation of the input power. As observed, a higher PAE can be achieved giving a higher input power until the 1 db gain impression point is reached. The extracted P in,ava, 1dB for both devices are tabulated and compared in Table 2.1. The presented results exhibit a stronger drain-bias-dependent P in,ava, 1dB variation as well 20

38 as a smaller P in,ava, 1dB value in the unpassivated device due to the larger power redistribution in harmonics as observed in Fig Unpassivated 40 Passivated P L,out,fun 0 P L,out,fun P L,out (dbm) P L,out,2nd P L,out (dbm) P L,out,3rd 60 P L,out,2nd P in,ava (dbm) 80 P L,out,3rd P in,ava (dbm) Figure 2.6: Measured output power P L,out versus available input power P in,ava at fundamental frequency (red), second (blue), and third (green) harmonic components. Four quiescent bias points are examined at (V GS, V DS )=(-3 V, 10V) (solid line), (-3 V, 15V) (dotted line), and (-3 V, 20V) (dashed line). Table 2.1: 1 db gain compression point P in,ava, 1dB versus drain bias voltage for both unpassivated and passivated GaN HEMTs Drain bias voltage 10 V 15 V 20 V Unpassivated 1.31 dbm 4.50 dbm 5.46 dbm Passivated dbm dbm dbm On the other hand, the average value of the dynamic drain current versus input power is of great interest as well since it is required to estimate other important device 21

39 30 Unpassivated 30 Passivated G L (db) & PAE (%) G L ( 3V,10V) ( 3V,15V) ( 3V,20V) PAE G L (db) & PAE (%) G L ( 3V,10V) ( 3V,15V) ( 3V,20V) PAE P (dbm) in,ava P (dbm) in,ava Figure 2.7: Measured power gain G L and power added efficiency P AE versus available input power P in,ava at fundamental frequency. Four quiescent bias points are examined at (V GS, V DS )=(-3 V, 10V) (solid line), (-3 V, 15V) (dotted line), and (-3 V, 20V) (dashed line). 44 Unpassivated 50 Passivated I DS,avg (ma) ( 3V,10V) ( 3V,15V) ( 3V,20V) I DS,avg (ma) ( 3V,10V) ( 3V,15V) ( 3V,20V) P in,ava (dbm) P in,ava (dbm) Figure 2.8: Measured average drain current I DS,avg versus available input power P in,ava. Four quiescent bias points are examined at (V GS, V DS )=(-3 V, 10V) (solid line), (-3 V, 15V) (dotted line), and (-3 V, 20V) (dashed line). 22

40 performances, such as PAE, power dissipation, device temperature as well as circuit reliability [5]. As a result, the measured average drain current I DS,avg versus available input power is presented in Fig Generally speaking, when the device is biased at a constant drain bias voltage, the average drain current will gradually increase with growing input power as illustrated in the case of the passivated device when V DS is equal to 10 V. However, as V DS increases, I DS,avg starts to deviate from its initial bias point even with a low P in,ava which suggests a different power-dependent nature of the dispersion effect at different drain bias voltage. In the unpassivated device, a notable power dependence of I DS,avg is observed. Also, I DS,avg begins to decrease even when a small input power is applied. However, the reduction in I DS,avg is smaller at higher drain bias voltage in contrast to the observations found in the passivated device. Such a discrepancy further suggests that different dispersion effects are encountered in the passivated and the unpassivated devices. With measured device output power and average drain current, it is easy to calculate the power dissipation from the power balance equation Eq. 2.8 as shown in Fig For both devices, the power dissipation reduces with increasing input power due to better PAE performance until the 1 db compression point is reached. While both the unpassivated and the passivated devices exhibit an increase of the power dissipation at higher drain bias voltage due to higher delivered output power, a lower power dissipation is observed in the unpassivated device due to a large reduction of the output power caused by more profound dispersion effects. From the above single tone characterization results, it becomes apparent that the unpassivated device suffers from lower output power, higher nonlinearities, lower power gain, and lower PAE while SiN passivation has been demonstrated to improve 23