Electromagnetic Compatibility in Industrial Equipment

Transcription

1 Electromagnetic Compatibility in Industrial Equipment Giorgio Spiazzi Department of Electronics & Informatics University of Padova - ITALY Fundamental Definitions Electromagnetic Compatibility The capability of electrical and electronic systems, equipment, and devices to operate in their intended electromagnetic environment within a defined margin of safety, at the specified level of performance, without suffering or causing unacceptable degradation as a result of electromagnetic interference 2

2 Fundamental Definitions Electromagnetic Interference (EMI) EMI is the process by which disturbing electromagnetic energy is transmitted from an electrical device to another via radiation and/or conducting paths In common terms, EMI refers particularly to RF signals, but it can occur in any frequency range starting from DC 3 Introduction Environmental electromagnetic pollution High-level electromagnetic disturbances can prevent electrical and electronic devices from operating properly Aspects to be considered emission of electromagnetic noises, and susceptibility to electromagnetic noises 4 2

3 Fundamental Definitions Susceptibility A relative measure of a device or a system propensity to be disturbed or damaged by EMI exposure to an incident field of signal Immunity A relative measure of a device or system ability to withstand EMI exposure while maintaining a predefined performance level 5 EMI Transmission Paths Source Channel Receiver Interference can be reduced acting on: Source (layout, filtering, shielding) Channel (layout) Receiver (layout, filtering, shielding) EMI is more conveniently suppressed at the source level in the design phase 6 3

4 EMI Problems Relevant EMI Components Only one third of the components affecting EMI are on the schematics Another third are parasitic elements within components The final third are created by PC board trace routing, and component mounting, placement, and even orientation 7 Resistivity of a Cylindric Conductor D W Uniform current distribution inside the conductor DC: 4 rdc = 2 σπ DW [ Ω /m] 8 (σ = conductivity) 4

5 Resistivity of a Cylindric Conductor D W δ Nonuniform current distribution inside the conductor due to skin effect AC: δ = µ 0 σπ f Skin depth (σ = conductivity) 9 r AC D W µ 0 πσ f [ Ω/m] Conductor Internal Inductance r W i_dc µ 0 = 8π [ H/m] µ i 0 4πrW πσ f / [ H m] The internal inductance plays no role at high frequency 0 5

6 r W External Inductance I s I s µ 0 cosh s e / π 2rW [ H m] Capacitance s Homogeneous medium: e c =µ ε 2 µ ε c = e π ε0 = cosh 2r s W [ F m] 0 0 / 6

7 Example Series impedance of two conductors AWG20 (26x34) at 2 mm distance (copper) External inductance: DC Resistance: l e 620 nh/m r DC 33 mω/m 3 Impedance per Unit Length Above some tens of KHz inductance is of major concern! [db] z () f r DC r () f r DC [MHz] 7

8 Resistors ) Wire wound 2) Film type 3) Composition R ideal L p > L p2 > L p3 R L p 5 Z( jω) = jωl C P R + + jω p RC p Resistor Parasitic Inductance Except for wire wound resistors, parasitic inductance is mainly due to lead length and separation S = 20mm PCB L = 8mm L p = 2.5nH 6 (C p = C pext +C loss = 0.06pF+(0. 0.5)pF) 8

9 Z(jω) dbω Example Bode plot of a MΩ resistor with L p =2nH, C p =0.5pF 00 f p =38kHz f 0 =2GHz [Hz] f p f 0 7 For high value resistors, parasitic capacitance is of major concern R s Capacitors L C R p R s = series resistance (ESR) R p = parallel resistance (dielectric losses) L = series inductance (ESL) 8 ( ω) Zj = Rs + jωl+ + R p jωr C p 9

10 Impedance of a Real Capacitor ω C Z(jω) R p + R s R s L R p C ω L R s 9 ω o = ω LC Inductors R p L C p R p = series resistance C p = distributed winding capacitance Rp + jωl Z( jω) = + jωr C ω p p 2 LC p 20 0

11 Impedance of a Real Inductor R p Z(jω) ω L C p L R p ωc p 2 ω o = LC p ω Inductors They can generate stray magnetic fields (air core and open magnetic core inductors are most likely to cause interference) They can pick up external magnetic fields For good high frequency behavior keep parasitic capacitance at minimum value (few separated turns can provide a higher impedance at high frequency than many turns closely spaced) 22

12 R L C RLC Network + u c u C (0)=V o u C V o αt ( t) = e sin( βt+ ϕ) Q 4 2 Q Z Z o = > R 2 o = L C α = ω o 2Q β = ω o 4Q 2 23 ω o = LC tg( ϕ) = 4Q 2 Ringing in a RLC Network In response to a perfect step input: Q = W.C. Overshoot = 6% Q = 2 W.C. Overshoot = 44% 24 The worst-case overshoot happens only if the step input generator can transmit significant energy at frequencies above the ringing frequency f o 2

13 Example: real step signal 90% 0% fmax = 2 T r T r (for a random gaussian pulse signal) If f max = f o actual overshoot is about one half of worst-case overshoot 25 Example Wire-wrap wire (AWG30) of length X = 2.25in, with heigth h = 0.2in above ground connecting a TTL driver (R out = 25Ω) to a typical load (C = 0pF) 0 AWG Wire diameter: d= 0 20 = 0. 0in 26 Q h L = X 4 9 ln = 50nH d Z o = = 283. fo = = R 2π LC MHz 3

14 Response to a Finite Rise-Time Step Input Signal.5 T r = ns T r = 2ns U U o in 0.5 T r = 4ns Time [ns] Example: TTL Logic Driving a Capacitive Load L +5V TTL output stage Q R c I L Load + Q 2 U gng U C - + L L L gng 28 4

15 U L Example: TTL Logic Driving a Capacitive Load L +5V I gnd U gnd t t TTL output stage Q R c I L Load + Q 2 U gng U C - + L L L gng 29 t di dt L U gnd = Lgnd = L gnd C L d 2 UL 2 dt EMI Common Sources Electronic circuits Diode recovery Switching components (SCR, IGBT, MOS) Driving current pulses Digital gates Magnetic components Transformers Inductors Circuit layout High dv/dt in long wires High di/dt in wide loops Mechanical switches relay (bounces, sparks, inrush currents) 30 5

16 Conducted and Radiated Signals Differential mode Common mode Earth Earth 3 Differential Mode Signals I I Ud= U U Id= 2, 2 2 i U d i 2 + E.U.T + U 2 U Earth 32 6

17 Common Mode Signals U U + U 2 c =, Ic = I + I 2 2 i i c U c + i 2 + U 2 U E.U.T Earth 33 Example of Conducted and Radiated EMI Through Power Supply Cables Boost rectifier Power Source i n i' i" 34 7

18 Measured Conducted Noise on Power Supply Cables Line Impedance Stabilization Network i dm Power Source LISN i cm 35 Example of Line Impedance Stabilization Network 50Ω/50µH R = 000 Ω C = µf C2 = 0. µf R2 = 50 Ω L = 50 µη R5 = R3 = 0Ω 36 8

19 EMI Filters Goal: to block conducted interferences close to their sources Power Source LISN EMI FILTER i dm i cm 37 Conducted Noise Common mode: parasitic capacitances between AC hot traces and chassis Differential mode: current ripples and spikes + i noise - v noise Switching devices can be seen as noise generators impressing noise voltages or injecting noise currents in the circuit Filters and parasitic elements must be taken into account in order to determine noise current and voltage amplitudes 38 9

20 Radiated Electric Field Parasitic capacities to free space Noise coupled to core through winding-core capacitances 39 AC hot traces (HIGH dv/dt) High dv/dt nodes in a Forward Converter V cc V DRIVE COM 20

21 Radiated Magnetic Fields Typical input and output switching loops 4 stray transformer and choke fields Transformer External Leakage Field Dipole Field Sec Pri Core Pri Sec Pri Sec MMF Common (worst) 42 2

22 Why High dv/dt and di/dt are a Problem? Spectrum Envelope of a Trapezoidal Periodic Signal x(t) A 43 τ d = τ T T T r = rise time 2Ad duty - cycle t dbµv f = π τ 2Af π f f 2 o = π t r 2Af π o 2 2 ftr f [MHz] Duty-Cycle Variations dbµv d 2 < d 2Ad 2Ad2 44 f o o π d fd π 2 f [MHz] 22

23 Rise and Fall Time Variations 2Ad dbµv τ r2 < τ r 45 πτ r πτ r 2 f [MHz] Repetition Frequency Variations f 02 > f 0 2Ad dbµv 46 f o πd f o2 πd f [MHz] NOTE: the distance between discrete harmonics nf 0 changes 23

24 Example: Drain-Source Voltage of a Switching Device U A = 600V Switch "off voltage t r = t f = 00ns rise and fall times f s = 00 khz Switching frequency C P = 30pF Parasitic capacitance between drain and ground C P i Cp DS = C du p = 80mA dt 47 Drain Voltage and Capacitor Current U DS 5µs 600V 0V i Cp 80mA 00nS 48 24

25 Drain-Source Voltage Spectrum [db] P - 20dB/dec - 40dB/dec f f 2 f 49 f P= 20log 2A τ 0 = 5.6dBµ V T = = 63kHz πτ s f = πt 2. f = 3 8MHz Capacitor Current Spectrum P [db] - 20dB/dec - 40dB/dec I = jωcu Cp p DS For f [db] < f < f 2 I Cp - 20dB/dec max f f f 2 f I = 4 U f C max Cp A s p 50 25

26 Common Mode Current C P i cm Power Source i p i n i dm i cm [db] i cm - 20dB/dec I Cp max 5 f f 2 f f 2 = π t r Example: Trapezoidal Waveform with Parasitic Oscillations u DS A 52 T S t 26

27 Neglecting parasitic oscillations dbv Spectrum Parasitic oscillations included A = 00V t r = 00ns t f = 00ns f s = 00 khz f o = 5 MHz d = f = π τ f = π 2 t r f [Hz] 54 Electrical Dimensions Physical dimensions of an electric or electronic circuit do not influence the capacity of a source to couple with a receiver Only electrical dimensions are important in determining the coupling efficiency between source and receiver Electromagnetic wave in a lossless medium: Wavelength: Propagation speed: λ = v = v f µε [ m] s m 27

28 Electrical Dimensions An electric or electronic circuit which is potentially a source of EMI is said to be electrically small if its physical dimension L is smaller than the wavelength of the electromagnetic wave generated L << λ Empirically: (L λ/0) 55 Electrical Dimensions For digital signals it is convenient to define the effective length of an electrical feature e, like a rising edge as: = vt e r = Tr D T r = rise time (0-90%) v = propagation speed D = delay [s/m] 56 28

29 Electrical Dimensions A digital trace or circuit can be considered a lumped circuit if its physical dimension L is much smaller than the effective length of rising and falling edges of digital signals L << e Empirically: (L e /6) 57 Example: Digital Signal Traveling on a PCB Inner Trace (ε r =4.5) V(t) T r =ns 0KH ECL Signal ns v = = µ ε vo ε r = [ m] s Effective length of rising edge: 58 e = vt r = 0.4m 29

30 V(t) 0 Signal Propagation Through Inner PCB Trace T r =ns 2 3 ns ns Elapsed Time V(t) m trace 59 Signal Propagation Through Inner PCB Trace V(t) T r =ns ns ns 2ns Elapsed Time V(t) m trace 60 30

31 Signal Propagation Through Inner PCB Trace V(t) T r =ns ns ns e = effective length e 2ns 3ns Elapsed Time V(t) m trace 6 Signal Propagation Through Inner PCB Trace V(t) T r =ns ns ns 2ns 3ns 4ns Elapsed Time e V(t) m trace 62 3

32 High dv/dt Coupling Mechanism: Parasitic Capacitance C 2 = parasitic capacitance between conductors and 2 Conductor 2 Conductor C 2 R U 2N U C 2G 63 Capacitive Coupling: Frequency Domain Equivalent circuit C 2 U C 2G R + U 2N Voltage noise U 2N induced in conductor 2 64 U2 U N = jω jω + RC ( 2 + C2G) C C2 + C 2 2G 32

33 Capacitive Coupling: Frequency Domain Voltage noise U 2N induced in conductor 2 U U 2N jωrc 2 C C C2 G 65 ω c = R C ( + C ) 2 2G ω Capacitive Coupling: Time Domain Assumptions: The coupled current flowing in C 2 is much smaller than the primary signal current in circuit. The coupled signal voltage in circuit 2 is smaller than the signal on circuit. A capacitor C 2 has a large impedance compared to the impedance to ground of circuit 2. The impedance to ground of circuit 2 is R 66 33

34 Capacitive Coupling: Time Domain U (t) 0 T r =ns U 2 3 ns Max dv/dt of driving waveform: du U = dt T Injected current in circuit 2: I C = C 2 U T r r 67 U 2N Crosstalk = = U RC T r 2 Capacitive Coupling Wherever there are two circuits, there is mutual capacitance. Voltages in one circuit create electric fields which affect the second circuit. A mutual capacitive coupling between two circuits and 2 is simply a parasitic capacitor C 2 connected from circuit to circuit 2. A mutual capacitance C 2 injects a current into circuit 2 proportional to the rate of change of voltage in circuit How can we reduce this coupling mechanism? 68 34

35 Cable Shielding Against E Field using a grounded shield U Conductor C S C 2S C SG Shield Conductor 2 R + U 2N - 69 Cable Shielding Against E Field Poor shield s ground connection : resonance between shield capacitance C SG and parasitic inductance L p of ground connection can actually increase noise voltage pick up by the inner conductor C 2S Shield C S L p 70 U C SG U 2N 0 35

36 Cable Shielding Against E Field Center conductor estending beyond shield Shield Conductor Conductor 2 C 2S C S U C 2 C 2G + - U 2N C SG 7 Cable Shielding Against E Field For good electric field shielding, it is necessary to minimize the length of the center conductor that extends beyond the shield and to provide a good ground on the shield 72 36

37 Electrostatic Shields in H.F. Transformers Unshielded Transformer Primary winding to core capacitance Core Secondary winding to core capacitance Primary to secondary winding capacitance 73 Electrostatic Shields in H.F. Transformers Shielded Transformer Alternative shield connection provided an input bypass capacitor is used Shield Noise current path 74 37

38 Electrostatic Shields in H.F. Transformers Incorrect shield connection Noise current path through large ground loops Common Mode Noise Minimization Minimize all sources of HF AC coupling to chassis Minimize AC hot traces on chassis side of PCB Use common traces to shield AC hot traces Avoid mointing AC hot cases on grounded heat sinks Tie power magnetic cores to AC quiet Minimize all coupling of HF AC from input to output Keep primary and secondary circuits well separated Nearest primary and secondary conductors should commons Use shields in isolation transformers 38

39 Minimizing Capacitive Coupling AC hot Out AC hot Out Return (AC quiet) Return (AC quiet) Increased spacing, hot to out Reduced AC hot trace area Guard or shield trace AC hot trace Output AC hot trace Output PCB PCB Ground plane 77 A ground plane reduces point-to-point PCB capacitive coupling Minimizing Capacitive Coupling AC hot trace Ground plane Chassis PCB Capacitive coupling AC hot trace Chassis PCB Ground plane A ground plane reduces PCB trace to chassis coupling 78 39

40 Shielding AC Hot Traces and Cases Shield 79 layout High di/dt Coupling Mechanism: Mutual Inductance 80 I (t) Loop loop 2 U 2N (t) + - Current in loop produces a pattern of magnetic field energy The total magnetic field strength over the area of loop 2 (magnetic flux) is a function of the distance, physical proportions, and relative orientation of the loops as well as of the current in loop 40

41 I (t) Loop Inductive Coupling U Faraday s law: dφ dt d 2 () t = = B 2N ds dt area2 H H 8 loop 2 U 2N (t) + - di U2 N() t = LM dt Mutual inductance: φ L M = I 2 Inductive Coupling: Frequency Domain I R 2 I 2 + L 2 U 2N - L M R Equivalent circuit 82 Voltage noise U 2N induced in conductor 2 U I 2 2N = jωl jωl U2N U = = R + R R I 2N T M I U jωl M 2N = I L 2 + jω R T 4

42 Inductive Coupling U i 2N jωl M L ML R T 2 R L T 2 ω 83 Inductive Coupling: Time Domain Assumptions: The induced voltage across L M is much smaller than the primary signal voltage in circuit. The coupled signal current in circuit 2 is smaller than the current in circuit. The secondary impedance is small compared to the impedance to ground of circuit 2. The impedance to ground of loop is R 84 42

43 Inductive Coupling: Time Domain I (t) 0 T r =ns U R 2 3 ns Max di/dt of driving waveform: di U = dt R T Injected voltage in circuit 2: U = L U 2N M RTr r 85 U 2N Crosstalk = = U LM R T r Inductive Coupling: Time Domain Example: U = V, T r = ns, L M = 0nH, R = 50 Ω Crosstalk = U U LM = R T 2 N = r Among high-speed digital circuits, mutual inductance is often a worse problem than mutual capacitance 43

44 Inductive Coupling Wherever there are two loops, there is mutual inductance. Current in one loop creates a magnetic field which affects the second loop. A mutual inductive coupling between two circuits and 2 is simply a parasitic mutual inductance L M between circuit and circuit 2. A mutual inductance L M injects a noise voltage into circuit 2 proportional to the rate of change of current in circuit 87 How can we reduce this coupling mechanism? Cable Shielding Against H Field using a shield grounded at both ends I R U R 2 I S Shield current can flow, thus inducing a second noise voltage on inner conductor 88 44

45 Magnetic Coupling Between Shield and Inner Conductor Noise voltage induced in inner conductor due to shield current L M2S U 2SN U SN L S I S R S I S = R S U SN + jωl S 89 U U 2SN SN = jωlm2s R + jωl S S U U 2SN SN = R L S S jω + jω Magnetic Coupling Between Shield and Inner Conductor U U 2SN SN U U 2SN SN jωlm2s = R + jωl S S U U 2SN SN = R L S S jω + jω 90 R ωc = L S S ω 45

46 Effect of Shield on Magnetic Coupling I R L MS + U SN R U + R 2 Noise voltage induced on shield 9 U SN = jωl I MS Effect of Shield on Magnetic Coupling I R U + 92 R 2 L M2 + + U 2N U R Noise voltage induced on conductor 2 by conductor 2N = jωlm2i 46

47 Effect of Shield on Magnetic Coupling I R L M2S + + R U + R 2 + U 2SN Noise voltage induced on conductor 2 by shield current 93 U 2SN = jωlm2sis Shielding Effectiveness Total noise voltage induced in shielded conductor 2 (L M2 = L MS ) U I 2N jωl M2 L M2 R L S S 94 R ωc = L S S ω U 2N = U 2N U jωl = L + jω R M2 S S 2SN I 47

48 I Shielding to Prevent Magnetic Radiation I I S U R L S L M I S R S U R I G I G 95 I S = + I RS jωl S The shield current, equal and opposite to the conductor current, produces a field which cancels that caused by the conductor Shielding Against Magnetic Fields Objective: reduce area of the receptor Loop I I U RL U R L I G =I I G =I I 96 U I S =I R L Shield grounded at both ends at ω > R S /L S 48

49 Shielding Against Magnetic Fields At low frequencies the shield should not be one of the signal conductor, due to the voltage drop on shield resistance R S caused by shield noise current I S + U in - L M R S + U in - L S I S 97 Ground point at different potentials U = R in I S S Cable Terminations An incorrect cable termination can destroy almost completely the shielding effectiveness for both emission and immunity. Only the use of a uniform 360 shield termination can preserve shield effectiveness Pigtail shield connection must be avoided 98 49

50 Shielding Against Magnetic Fields 99 At high frequencies a coaxial cable contains three isolated conductors: the center conductor the inner surface of the shield conductor the outer surface of the shield conductor The inner and outer surfaces of the shield are isolated from each other by skin effect Noise current flows on the outside and there is no common impedance 3 Common Impedance Coupling Path ( ) U = Z + Z + Z I + Z I s s L C C 2 Z s I Noise Voltage U s Z L Z C + U s2 Z s2 I 2 Z L2 00 How to reduce this coupling mechanism? 50

51 Signal ground (low level circuits) Grounding.using a good ground system Noisy ground (relays, motors, high-power circuits) Hardware ground (chassis, racks, cabinets) 0V 0 Keep separated to avoid noise coupling Chassis (Hardware) Ground Also often a safety ground Includes metallic chassis, framework, etc. Often tied to a true earth As a rule, not used as a power return conductor (except in some low-voltage DC applications, like autos) Not good as ground plane due to seams, joints and openings Also common mode conducted noise return 02 5

52 Circuit Ground Usually refers to circuit common or signal ground May be connected to chassis ground, but is best considered as separate and isolated during design Often the negative rail in DC systems, but the positive voltage rail may also be used AC Quiet Rail Also known as AC LOW or RF LOW Used for local HF bypassing in EMI control Typically one of the DC or LF AC supply rails Not necessarely the same as circuit common 03 Circuit Ground Equipotential point or plane that serves as a reference potential for a circuit or system 04 Does not exist in practice because any conductor have a finite impedance, both resistive and inductive. Inductance is the major problem at high frequency Two physically separated ground points are seldom at the same potential 52

53 Circuit Ground A low-impedance path for current to return to the source (Ott) Key point: To identify the actual current return path for each circuit in order to: 05 ensure separated return paths for different circuits (thus avoiding coupling through common impedances) control actual current loop area (minimize magnetic coupling) Ground Plane A large metallic area (typ. on a PC board), which serves as: Magnetic field reduction through image currents Electrostatic faraday shield Circuit common / power return AC quiet rail Thermal Heat spreader Printed circuit board stiffening Typically not connected to chassis ground, but may be in some cases (usually undesirable for EMI) 06 53

54 Ground Plane as Return Current Conductor A - Signal path B - Shortest low-frequency return path (smaller resistance) C - Shortest high-frequency return path (smaller inductance) U ac A C B Load Different current components follow different return paths Ground plane 07 Actual current path is that with lowest impedance Voltage drop on ground paths can be significant for high amplitude and high-frequency currents Ground planes are preferable in high frequency applications Magnetic Field Reduction with a Ground Plane Effect much higher behind ground plane front: reduction due to image currents (quadrapole effect) rear: eddy current field cancellation Ground plane slits and slots under conductors greatly reduce its effectiveness Thin solid ground plane better than thick planes with slits or slots 08 54

55 Monopole field: Field intensity /R 2 Far Field Intensity Dipole field: Field intensity /R 3 Quadrapole field: Field intensity /R 4 09 Ground Plane Effect E and H fields distribute themselves as if a mirror image conductor existed at the opposite side of the ground plane, with an equal current in the opposite direction Ground plane 0 Transmission line impedance Z o is half of imaged line impedance 55

56 Magnetic Field Reduction with a Ground Plane PC trace Image current Solid conductor ground plane Counter-flowing image currents caused by a ground plane create a quadrapole magnetic field Imperfections in the Ground Plane PC trace on the other side of the PC board 2 An open path in the ground plane creates a slot antenna A break in the ground plane further reduces image current effect 56

57 Imperfections in the Ground Plane Return current path from A and B (A) (B) The high loop area increases inductance and crosstalk with adjacent traces 3 Single-Point Grounding Series connection # #2 #3 Z Z 2 Z 3 I +I 2 +I 3 I2 +I 3 I 3 0V Cross-coupling due to common impedances in the ground lead 4 The most sensitive circuit must be closest to the physical ground point (#) 57

58 Single-Point Grounding Parallel connection # #2 #3 Z Z 2 Z 3 0V No coupling at low frequencies, but possible capacitive and magnetic coupling at high frequencies 5 High ground connection impedance due to possible long connections Multiple Ground Points # #2 #3 Z Z 2 Z 3 0V A very low-impedance ground plane is required (otherwise it behaves like a single point series connection) 6 The connection between each circuit and the ground plane must be kept as short as possible to reduce its impedance 58

59 Ground Systems When different types of circuits are present on the same board, group them in a appropriate manner and keep their ground connections separated Analog Noisy Digital 7 Ground Plane Separation Return currents of each circuit remain separated Subcircuit GND GND2 Subcircuit

60 EMI Filters How to block any unintended signal (EMI) on conductors (both signal conductors and power conductors), and. Power Source LISN i dm i cm 9 EMI Filters how to preserve the shielding effectiveness of an enclosure by preventing electromagnetic energy from entering or leaving the enclosure Antennas Use EMI Filters 20 60

61 Input Filter for Power Line Conducted Emissions The filters must be transparent to the power supply (AC line or DC) high component polarization Line and load impedances are not well defined The filters are required to operate in a very broad frequency range (filter components change their characteristics possible resonances) 2 22 CM and DM Filters Each power line filter is composed by a Differential Mode (DM) section and a Common Mode (CM) section DM section must be transparent to the power frequency (DC or AC line) lower limit on filter corner frequency and high component polarization CM section does not have these limitations (no useful signal to pass). However the maximun CM capacitor values in AC power line application are limited by leakage current to earth (safety reasons) 6

62 Example of CM and DM Filter C 3 C L CM C 2 C 4 Common Mode Differential Mode L CM C 3 +C 4 C L d C 2 +C 3 //C 4 23 Filter Configuration Key point: maximum impedance mismatch at input and output filter ports Z g U g C L Z L Z g U g L C Z L Z g = high, Z L = low Z g = low, Z L = high Z g U g C L C Z L Z g U g L C L Z L 24 Z g = high, Z L = high Z g = low, Z L = low 62

63 CM Inductor High inductance value through the use of a high permeability core low end-to-end parasitic capacitance so as to increase self resonance frequency 234 N N 234 N N N 25 Low end-end capacity solenoidal choke windings Low end-end capacity (used in high current RF chokes) CM Inductor End-end capacity can be dominated by winding-core-winding capacity S F S F Core Common 26 Low end-end filter choke capacity requires a grounded core to the filter circuit common or AC quiet rail 63

64 CM Inductor For capacity grounding of toroidal cores a wrap of copper tape is placed around the core Core 27 Effect of a High Contact Impedance Filter metallic enclosure C L C 2 U Noise Load Contact impedance 28 A high contact impedance of the filter enclosure nullify its filtering properties (C and C 2 bypass the series inductor) 64

65 Commercial Filter Installation Filter Load Load Filter Correct installation Incorrect installation: noise couples directly to input terminals 29 Commercial Filter Installation Incorrect installations Filter Possible input-output coupling Filter Ground Input and output filter connection must be separated to avoid crosstalk 30 65

66 Commercial Filter Installation Auxiliary shield Auxiliary shield Filter Filter 3 Insertion Loss Z + + S U Filtro U Z L g 2 - IL = 0 log 0 P P 2 2f = 20 log 0 U U 2 2f P 2 = load power without filter 32 P 2f = load power with filter 66

67 Insertion Loss The filter insertion loss depends on source and load impedances Commercial EMI filters are characterized by insertion loss measured with 50Ω source and load impedances 33 C.M. and D.M. Mode Filter Arrangement for DC and 20AC Cores bypassed Tied to internal AC quiet HI LO C DM G C CM L DM Closed to input terminals L CM Closed to circuit C DM C CM Switching unit Separate grounds 34 67

68 Poor Filter Layout Large loop areas (noise pickup) HI LO G Cores not bypassed L DM C CM L CM Noise Large loop areas (noise pickup) C DM C CM Common ground with high L Switching unit 35 Radiated Noise How we can prevent electromagnetic energy from leaving an enclosure, thus avoiding interferences with external apparatus (limiting emission)?... Noise source Shield 36 68

69 Radiated Noise. or how we can prevent electromagnetic energy from entering an enclosure, thus avoiding interferences with external apparatus (limiting susceptibility)? Antenna 37 Use a Shield Shielding Outline Analysis of shielding mechanism of an ideal homogeneus and infinite shield Absorption Reflection Effect of shield imperfections 38 69

70 Shielding Effectiveness Shielding: reduction of magnetic and/or electric filed strength caused by the shield Circuit theory point of view: Opposing field Conducting material 39 Incident magnetic field Induced current in conductor Shielding Effectiveness Field theory point of view: Ê S = 20log Ê Shielding effectiveness: Ĥ S = 20log Ĥ o t o t [ db] [ db] E o, H o : incident field strenght E t, H t : field strenght of the transmitted wave as it emerges from the shield 40 70

71 Shielding Effectiveness Two aspects must be considered: the shielding effectiveness of the shield material itself the shielding effectiveness due to discontinuities and holes in the shield We first consider the shielding effectiveness of a plane sheet of conducting material. Then the effect of discontinuities and holes is taken into account 4 The overall shielding effectiveness is dominated by discontinuities and holes in the shield Shielding Low-Frequency Magnetic Field Providing a low-reluctance magnetic path to divert the field around the circuit being protected Magnetic field Shield of magnetic material Shielded region 42 7

72 Shielding High-Frequency Magnetic Field Using short circuit loops to generate an opposing magnetic field Example: transformer Core Winding Core Conducting shield acts as short circuit loop (reduces leakage flux) 43 Apertures The intrinsic shielding effectiveness of a material is of less concern than the leakage through seams, joints, and holes The amount of leakage from a shield discontinuity depends on: the maximum linear dimension (not area) of the opening the wave impedance the frequency of the source 44 72

73 Effect of Shield Discontinuity on Magnetically Induced Shield Current Induced shield current Rectangular slot 45 Apertures A large number of small holes produce less leakage than a large hole of the same total area A circolar hole produces less leakage than a rectangular one of the same area 46 73

74 How to Reduce H Field Emission from Ring Core Inductors and Wide Current Loops Ring core Inductor Snubber component layout 47 How to Reduce Transformer External Leakage Field Dipole Field Quadrapole Field Sec Pri Core Pri Sec Pri Sec Pri Core Pri Sec Pri Minimum external field Pri Sec Pri Sec Pri MMF Common (worst) MMF 48 74

75 How to Reduce H Field Emission from Printed Circuit Power Conductors + Best, but usually not practical Very good, sometimes practical How to Reduce H Field Emission from Printed Circuit Power Conductors Quite good, usually practical for standard practice Image currents Ground plane Acceptable, due to quadrapole field from ground plane Image currents Ground plane 75

76 How to Reduce H Field Emission from Printed Circuit Power Conductors Poor, large dipole field and high AC losses due to current concentration on adjacent edges + - Unacceptable, very large dipole field (lower AC losses than the previous pattern) How to Reduce Parasitics Connection of Capacitors Mutual inductance coupling. The same L m coupling occurs with shorted, isolated PC traces of the same geometry (typically 2-20 nh) I in L m Coupling L m Coupling 52 76

77 How to Reduce Parasitics Connection of Capacitors Some ESL Some L m Close spacing 53 Preferred alternative How to Reduce Parasitics Feed-through capacitors Chassis Schematic representation Equivalent circuit 54 The only lead inductance present is in series with the signal and transform the feedthrough capacitor into a low-pass T filter 77

78 Paralleling Capacitors No one capacitor can provide satisfactory performance over the entire range from low to high frequencies Two or more capacitors of different types and values are ofted used in parallel to provide a low impedance in a broad frequency range Be aware: paralleling of capacitors can cause resonances 55 Carefull layout to reduce parasitic inductances 56 General Layout Considerations for Switching Power Converters In general, the power, drive, and logic physical layout should resemble a neat schematic Keep isolated circuits physically separated Keep noisy power circuits away from logic and low-voltage control circuitry Keep power switching circuits away from filtered inputs and outputs Place drivers close to switches and away from logic Parallel discrete or dual diodes with caution: different T rr can excite HF oscillations 78

79 General Layout Considerations for Switching Power Converters 2 Minimize component lead inductances surface mount components are preferred radial leaded components should be mounted normal to the PCB (i.e. standing up) axial leaded components should be mounted parallel to the PCB (i.e. laying down) keep leads as short as possible (ESL is proportional to lead length keep snubber and clamp loops as small and close to snubbed devices as possible 57 Linear Power Path Minimize noise coupling from switching circuits into input and outputs Input Input filter Switching & rectification Output filter Output Drive Sensing Aux Control Iso. Aux 58 79

80 Folded Power Path Still good layout 59 Input Aux Output Input filter Control Iso. Sensing Output filter Drive Switching & rectification Folded Power Path 2 Less desirable layout Input Input filter Switching & rectification Drive Control Iso. Aux Output Output filter Sensing 60 80

81 6 Summary Control of emission/susceptibility requires a careful system configuration, starting from PCB layout (many parasitic components on your PCB are created by an inaccurate layout) A good ground system minimizes the noise voltage from ground currents flowing through a common impedance and reduces radiated emission through image currents Shielding can effectively reduce EMI propagation but its effectiveness is dominated by discontinuities and holes in the shield (remember: the amount of leakage from a shield discontinuity depends on the maximum linear dimension, not area, of the opening) Input filters are useful in reducing conducted and radiated EMI produced by EUT (attention: If not properly damped, an input filter can cause instability and system degradation when used in switching power supplies) 8