Structure-Dependent Dielectric Constant in Thin Laminate Substrates

Transcription

1 Structure-Dependent Dielectric Constant in Thin Laminate Substrates by Hyun-Tai Kim, Kai Liu*, Yong-Taek Lee, Gwang Kim, Billy Ahn and Robert C. Frye STATS ChipPAC, Ltd. San Ami-ri Bubal-eup Ichon-si Kyonggi-do Korea Tel: *STATS ChipPAC, Inc West Greentree, Ste 117 Tempe, Arizona 8528, USA Tel: Copyright RF Design Consulting, LLC 33 B Carlton Avenue Piscataway, NJ 0885 USA Tel: bob@rfdesignconsulting.com Reprinted from 2011 Electronic Components and Technology Conference (ECTC) Proceedings. The material is posted here by permission of the IEEE. Such permission of the IEEE does not in any way imply IEEE endorsement of any STATS ChipPAC Ltd s products or services. Internal or personal use of this material is permitted, however, permission to reprint/republish this material for advertising or promotional purposes or for creating new collective works for resale or distribution must be obtained from the IEEE by writing to pubs-permission@ieee.org. By choosing to view this document, you agree to all provisions of the copyright laws protecting it.

2 Structure-Dependent Dielectric Constant in Thin Laminate Substrates Hyun-Tai Kim, Kai Liu*, Yong-Taek Lee, Gwang Kim, Billy Ahn and Robert C. Frye STATS ChipPAC, Ltd. San Ami-ri Bubal-eup Ichon-si Kyonggi-do Korea Tel: * STATS ChipPAC, Inc West Greentree, Suite 117 Tempe, Arizona 8528, USA Tel: RF Design Consulting, LLC 33 B Carlton Avenue Piscataway, NJ 0885 USA Tel: bob@rfdesignconsulting.com Abstract This paper describes an experimental method for extracting the frequency-dependent of board material. The method is applied to thin laminate substrates used for SiP substrates at frequencies up to 10GHz. Cross-section measurements are used to determine layer thicknesses and etched linewidths in the experimental samples. The extraction method uses a combination of electromagnetic simulation and measurement to extract the from propagation delay measurements. Characteristics extracted from an example line are used to model a variety of other transmission line geometries with good agreement. For the example substrate used in this investigation, the was significantly lower than expected compared with the manufacturer s specifications. A possible reason for this is that in very thin boards, it is not always possible to achieve the same ratio of resin to reinforcement as in thicker boards with multiple layers of glass cloth. Consequently, the board composition in thin packages may not precisely match the average composition of more conventional, thicker boards. The delay-based results are compared with more conventional capacitance-based methods. The results in both cases are similar, but the capacitance method requires much more accurate determination of the layer thickness, which is difficult to achieve in thin laminates. Comparisons of odd- and even-mode propagation delays in differential transmission lines show no conclusive evidence of appreciable anisotropy in the board material studied. Modal dispersion is only weakly dependent on anisotropy, making experimental determination by this method difficult. Introduction Package thickness is an especially important system requirement in portable consumer electronics. Laminates for typical SiP substrates use very thin (less than 100 micron) layers. Modern RF wireless applications have performance requirements in the multi-ghz frequency ranges. Traditional technological approaches to such applications have used specialized laminate materials that offer very good electrical performance at these frequencies. However, the cost and size constraints on SiP substrate materials typically rule out the use of such specialized materials in these applications. Because the packages are small, RF losses in more conventional laminate materials are usually acceptable. This paper examines the use of low-cost, thin laminate materials for RF wireless applications. In particular it looks at the electrical characteristics of these materials as they apply to the propagation of multi-ghz signals in SiP substrates. Materials Laminate materials for packages are most often used in applications other than RF wireless. Key criteria for their development and selection are mechanical (stiffness and flatness) thermal (glass transition temperature) and ecological. Manufacturers data sheets for the RF electrical characteristics often have very limited information, especially at frequencies of interest for wireless applications. Laminates are a composite material, typically made up of an inorganic reinforcing framework material impregnated by an organic resin. Within a particular class of material, they are available in a wide variety of thicknesses. In a typical laminate structure, and in the particular laminate used in this investigation, the reinforcing framework is a woven glass cloth [1]. In such a structure, the glass generally has a higher dielectric constant than the resin, and the composite material has an intermediate value. On a fine scale, the dielectric properties of low-cost laminates are inhomogeneous. This has been shown to be a particular problem in thin layers. To maintain a desired value of characteristic impedance in transmission lines, it is necessary to scale the dimensions of the lines in proportion to the dielectric thickness. Consequently, thin laminate packages require narrow line dimensions. For the very thin laminates used in SiP applications, the line dimensions are comparable to the size of the texture in the woven glass reinforcing cloth, so the inhomogeneity in the electrical properties of the laminate may become apparent. One well-documented manifestation of this inhomogeneity is the so-called weave effect [2,3]. Transmission lines that are aligned with the weave of the cloth may lie over glass-rich (i.e. higher ) portions of the laminate, or alternatively they may lie over more resin-rich (i.e. lower dielectric constant) regions. Studies of this effect have found local variations in the propagation velocity on the order of % (i.e. about 8% variation in ) depending on the orientation of narrow lines with respect to the underlying weave. This effect is especially of concern in the design of differential transmission lines, where velocity skew between adjacent lines is undesirable. Another effect of the inhomogeneity is anisotropy in the dielectric properties. The woven glass cloth lies in the plane of the board. Electric fields normal to the plane of the board pass through either the glass-rich higher-capacitance regions of cloth or through the resin-rich lower-capacitance regions in the holes of weave. For a normal field these regions are seen in parallel. For tangential fields, however, these inhomogeneous regions are seen in series, in which case the overall capacitance is dominated more by the lower /11/$ IEEE Electronic Components and Technology Conference

3 of the resin. Studies have found the anisotropy in low-cost board materials to be as high as 10% [,5]. A third and less well-documented effect of the board composition is that the dielectric properties show thickness dependence. Laminate manufacturers typically choose from a variety of standard cloth thicknesses, composition and weave configurations. In thicker boards made up of several layers of cloth, it is usually possible to maintain the ratio of glass to resin so that the of the board is kept close to its nominal values. However, in very thin boards this is not always possible. For example, for layer thicknesses of either 50 or 60 microns, using standard glass cloth compositions it is only possible to accommodate a single layer of 35μm-thick glass cloth. So the difference between these two thicknesses is entirely made up by adding more resin. Consequently, greater variation in the composite can be expected in thinner boards. In thicker boards, multiple layers of different types of cloth are used to fine-tune the composition with greater accuracy. Experimental The overall layout of the laminate test board is shown in Figure 1. Figure 2: Test board nominal structure A common problem in the fabrication of laminate substrates is linewidth control. (This is especially true for onetime fabrication of experimental structures, for which only nominal fabrication processes are used and linewidth control parameters have not been adjusted.) Physical cross-section measurements were done to determine the dielectric layer thicknesses and metal linewidths in the fabricated structures. The average values extracted from these measurements were used in subsequent electromagnetic simulations. Table I shows the results of cross-section measurements for the transmission line structures. The electrical test structures examined in this study were all fabricated in the upper prepreg layer. The lower metal-2 copper layer served as the ground plane and the test structures shown in Figure 1 were patterned in the top metal-1 copper layer. Several of the typical characteristics of the prepreg layers as specified by the manufacturer are listed in Table II. TABLE II MANUFACTURER S SPECIFICATIONS PROPERTY VALUE Figure 1: Overall laminate test board layout The test devices on the board are designed to be probed using 150µm pitch GSG probes. The test board has a variety of single ended microstrip transmission lines with widths ranging from 50 to 150µm, and with lengths of 10 and 20mm. Differential microstrip transmission lines with a variety of line widths, and separations and two parallel-plate capacitors are also included in this test board. Parallel-plate capacitors are formed between the top metal and the next underlying layer, and can be used to characterize the prepreg layer capacitance characteristics. The board is a -layer construction. The stack-up and nominal layer thicknesses are shown in Figure 2. TABLE I LAYER THICKNESSES AND LINE WIDTHS MEASURED FROM CROSS-SECTIONS FOR THE TRANSMISSION LINES (MICRONS) LINE WIDTH M1/M M2/M3 PPG CORE Material type CCL-HL-832-HS Manufacturer Mitsubishi Gas Chemical Thickness (nominal) 0 microns ε r 1MHz 5.1 ε r 1GHz.7 ε r 5GHz.6 ε r 10GHz.5 ε r 20GHz.5 Simulation and Dielectric Constant Extraction The electromagnetic simulations were done using EMX [6], with layer thicknesses and etched line widths determined from the cross sections. The materials used in the fabrication of laminated package substrates are generally non-magnetic i.e. their permeability is the same as that of free space. An important consequence of this is that the inductive part of their electromagnetic behavior is entirely determined by the geometry of the conductors. In particular, it is independent of the. So, within the tolerance of the crosssection measurements there are no adjustable parameters in this part of the simulation. 050ⅹ / / ⅹ / / ⅹ / / ⅹ / / ⅹ / /

4 Figure 3: The conventional ladder-network model of a transmission line The conventional ladder-network model of a transmission line, shown above, consists of a sequence of infinitesimal series impedance per unit length elements Z S and parallel admittance per unit length elements Y P. The characteristic impedance is given by Z Z S 0 (1) YP And the propagation coefficient is given by 1 Z S Y (2) P If the lines are not too lossy, the inductance dominates Z S and the capacitance dominates Y P. For non-magnetic materials, the inductive part of the series impedance is entirely determined by the dimensions of the conductors. So, given the cross-sectional measurement results for these lines, there are no adjustable parameters in this part of the simulation. The parallel admittance, on the other hand, is mainly determined by the capacitive part of the line s behavior (i.e. the electric field.) This is directly dependent on the, which is the parameter of interest in this study. It can be extracted from either the characteristic impedance or from the propagation constant. From a measurement perspective, the latter is preferable since it determines the propagation delay through the transmission line. This parameter can be measured with a very high degree of precision. The method for extracting the from the transmission line method consists of first simulating the line characteristics assuming several different values of. Then, the measured propagation delay is plotted as a function of frequency alongside the various simulations. This is shown in Figure. The simulated curves in Figure correspond to dielectric constant values of.0, and 3.6. The propagation delay is calculated from P arg( S 21 ). (3) Figure : The various simulated and measured propagation delay In general, Eqn. (3) is a valid basis for comparison provided both the simulation and measurement are evaluated under the same operating conditions. In practice, the measured S- parameters are typically normalized to 50Ω port impedances. For evaluation, it is preferable to renormalize the results to a value close to the characteristic impedance of the line. This minimizes line resonances. So, the procedure used to generate the result shown in Figure was to first determine the characteristic impedance of the measured line from Z MEAS re[ Z 11 / Y11 ]. () All measured and simulated results shown in the example in Figure are normalized to the low frequency measured impedance of 35Ω. Using this type of plot, at any frequency the fitted value of the can be found by interpolating between simulation curves. Figure 5 shows the frequency-dependent extracted from three samples. In the figure, the solid line is a 3 rd -order least-squares fit to the experimental data. Figure 6 shows a comparison of simulation and measurement using this frequency-dependent fitted dielectric constant. The parameters shown in the plot are the open circuit input admittance (magnitude and phase) and the short circuit input impedance (magnitude and phase). It can be seen here that the agreement between simulation and measurement is excellent using the fitted Figure 5: The frequency-dependent 1719

5 plotted against simulated values assuming different dielectric constants. (a) Figure 8: The simulated capacitance assuming different s and measured capacitance (b) Figure 6: (a) Open circuit input admittance (b) Short circuit input impedance These simulation results are based on the extracted from measurements of a 150-micron wide transmission line. Applying the same model for the dielectric behavior to other line types of equivalent length but different line widths results in somewhat less accurate predictions of the propagation delay. The delay error (measured at 5GHz) for different line widths is shown in Figure Using the same method of interpolation at each frequency, a frequency-dependent can be extracted that best matches the capacitance data. The derived from capacitance measurement for three different samples is shown in Figure simulation error (ps) line width ( m) Figure 7: The delay error for different line widths Note that for the 150-micron lines used to extract the dielectric constant, the agreement is quite good. Average error is less that 0.01ps, or about 0.2% of the total delay. For other line widths, the worst-case error is about 3.5ps, which is about 2% of the total delay for that line. Capacitance measurement comparison The same approach can be used to extract the dielectric constant from a large parallel-plate test capacitor included on the substrate. An example measurement is shown in Figure 8, Figure 9: The derived from capacitance measurement The capacitance simulations shown in Figure 8 were based on cross-section measurements of the capacitors. Capacitance-based extraction of the is not as good as delay-based extraction because the capacitance is directly related to the prepreg layer thickness. Consequently, any error in the thickness measurement is directly reflected in the simulation. In the boards used in this study, as with most laminate substrates, the metal/insulator interface is intentionally given some textured roughness to improve the adhesion between the two. This makes precise determination of the prepreg thickness problematic. 1720

6 (a) (b) Figure 10: Example cross-section micrographs comparing the structure of thin (a) and thick (b) laminates. Figure 10 shows example micrographs comparing a cross section for one of the thin laminate structures examined in this study with that of a more conventional, thicker board. (Note that the magnification in these two micrographs is not the same.) The glass reinforcing fibers can be seen in crosssection as the dark circular regions within the laminate. These fibers are concentrated in the oval-shaped bundles. The width of the bundle, depending on glass cloth type, is μm. The cross section for the thin laminate demonstrates the problem of accurately specifying the layer thickness in this case. The boundary between the dielectric and metal shows small-scale roughness and larger-scale undulations. These local variations make it difficult to determine the average thickness with a high degree of relative accuracy. For the transmission lines the delay is directly related to the but it is not especially sensitive to board thickness. Thicker boards have lower capacitance per unit length, but they also have higher inductance per unit length. These two effects tend to offset one another in the delay. Consequently, the delay-based determination of dielectric constant is much less sensitive to thickness than the capacitance-based results. This is especially true for thin boards like the ones examined in this study. The extracted from the capacitance can be seen to be similar in value to the extracted from the transmission line delay. In particular, the is found to fall with increasing frequency. The capacitance-based results show greater spread in the value of, especially at lower frequencies. This is probably a result of the sensitivity of these results to the board thickness, which varies somewhat from sample to sample. This same variation is present in the transmission lines, but as discussed above the delay-based results are not as sensitive to board thickness. Differential transmission lines Recently, there has been much interest in the anisotropic behavior of circuit board materials []. The degree of anisotropy in the board material can be obtained by comparing the odd- and even-mode delay characteristics of differential transmission lines. In the even mode, the local voltage on the two adjacent lines is the same. In this configuration, the electric field has a strong vertical component, normal to the plain of the board. In the odd mode, however, the local voltage on the lines is of the opposite polarity. Consequently, there is a stronger lateral field between the two, in the plane of the board. The simulation-based method described above can be used in both cases to extract the frequency-dependent dielectric constant. It can be seen in Figure 1 that the differential transmission lines in this study were designed for measurement using GSG probes. The measurements for these lines were performed using a -port VNA. Consequently, the problems of resonant coupling between the modes described in [] were not a concern in this case EVEN MODE ODD MODE 3.75 Figure 11: Extracted odd- and even-mode for the 75μm linewidth, 125μm space differential transmission line. Figure 11 shows a comparison of the even-mode and oddmode s extracted using the simulationbased method. The difference between the two results is nowhere greater than about 0.5%, and averaged over all frequencies the difference is nearly zero. Table III shows a comparison of the measured and simulated ratio of even-mode to odd-mode delay. The values listed in the table are the average 1721

7 over frequency from 100MHz to 10GHz. Both simulated and measured ratios show small (±0.05%) variations with frequency, from either measurement uncertainty or numerical noise. CONDITION TABLE III DELAY RATIO (EVEN/ODD) VALUE Measurement , , Simulated, ε r= Simulated, ε r= Simulated, ε r= Within the accuracy of the measurement and the sampleto-sample variation, there is no clear experimental evidence to indicate significant anisotropy in these boards. This may partially reflect the fact, as pointed out in [], that the modal dispersion in differential lines like these is not very sensitive to the anisotropy. In [], for example, in differential lines on FR a 10% difference between in-plane and vertical dielectric constant was proposed to account for an observed 1% difference in delay, approximately 2% difference in effective dielectric constant. In these measurements, however, the differences between effective in even and odd modes are at most 0.5%. The observed anisotropy is at least four times smaller than the values measured for FR, and is negligible compared with other sources of experimental error. Discussion and Conclusions A simulation-based method has been presented that can be used to determine the frequency-dependent. This method is general and can be applied to a variety of structures. It is especially accurate when applied to characteristics that depend directly on, such as transmission-line delay or capacitance. A key feature of this methodology is that it requires independent experimental methods, (e.g. cross-sections) to accurately determine the conductor dimensions. With this information, the dielectric constant is the only undetermined parameter in the simulations. For the low-cost, thin substrate material examined in this study, the derived from transmission line measurements from dc to 10GHz showed a 7% decrease. This is similar to the manufacturer s specifications showing a 10% decrease. In contrast, however, the overall measured value was nearly 17% lower than the specifications. This result was surprising, and suggests that the composition of very thin boards may be significantly different from that of more typical boards. As the cross-section in Figure 9a shows, in such thin boards it is only possible to include a single layer of glass reinforcement cloth. Consequently, it is not possible to adjust the ratio of glass to resin in thin boards as precisely as in thicker boards like the example in Figure 9b. This may account for the observed discrepancy. If this supposition is correct, it demonstrates the need to characterize thin board materials for sensitive RF applications. Parallel-plate capacitor measurements showed similar results. However, the capacitance is sensitive to the board thickness, and the accuracy of the capacitance-based measurement is limited by the accuracy of the thickness measurement. The relative error in thickness measurements is a particular problem for thin laminates. Delay-based results, on the other hand, are primarily dependent on line length and dielectric constant, and are relatively insensitive to layer thickness. For this reason, we have more confidence in these results. The same delay-based extraction of the for differential transmission lines also showed similar values for the. Interestingly, the s extracted from even and odd propagation modes were not appreciably different. The average measured delay ratio for the three samples examined fell within 0.3% of the expected value in all cases, assuming an isotropic. This is less variation than observed in FR samples []. References 1: G. Brist, B. Horine and G. Long, Woven Glass Reinforcement Patterns, Printed Circuit Design and Manufacture, 28-33, November : S. McMorrow and C. Heard, The Impact of PCB Laminate Weave on the Electrical Performance of Differential Sinaling at Multi-GigaBit Data Rates, DesignCon : C. Herrick, T. Buck and R. Ding, Bounding the Effect of Glass Weave through Simulation, DesignCon : J. C. Rautio and S. Arvas, Measurement of Planar Substrate Uniaxial Anisotropy, IEEE Trans. Microwave Theory and Techniques, 57, , Oct : J. Baker-Jarvis, B. Riddle and M. D. Janezic, Dielectric and Magnetic Properties of Printed Wiring Boards and Other Substrate Materials, NIST Technical Note 512, March : S. Kapur and D. Long, Large-Scale Full-Wave Simulation, DAC 200, San Diego June 7-11,