LINEAR MOTION is an essential requirement in many

Transcription

1 4346 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 61, NO. 8, AUGUST 2014 Transverse-Flux-Type Cylindrical Linear Synchronous Motor Using Generic Armature Cores for Rotary Machinery Jung-Seob Shin, Member, IEEE, Ryuji Watanabe, Student Member, IEEE, Takafumi Koseki, Member, IEEE, and Houng-Joong Kim Abstract This paper presents the design and analysis of a transverse-flux-type cylindrical linear synchronous motor using generic armature cores for rotary machinery that can address the problem of complex structures in conventional transverse-fluxtype topologies. First, the operational principle and structural advantages of the proposed model are explained. The thrust density and cogging force are investigated during the initial design stage using an application in which large thrust density and low cogging force are required. The proposed model is both theoretically and numerically designed by using a magnetic-circuit method and a 3-D finite-element method, respectively. Finally, the results and efficacy of our structural concept are experimentally validated. Index Terms Cogging force reduction, finite-element (FE) method (FEM), permanent-magnet linear synchronous motor (PMLSM), thrust design, transverse-flux-type machine. l g l gc τ p a z B g B r A m A g μ rm l m N E rms p I H c d a l c K cp S c NOMENCLATURE Air-gap length. Air-gap length considering the Carter coefficient. Pole pitch. Half length of the field magnet. Distance in the moving direction. Air-gap flux density under no-load conditions. Remanent flux density of the field magnet. Dimension of a magnet. Dimension of the air-gap area. Relative recoil permeability of magnet. Magnet length in the magnetization direction. Number of winding turns on a salient pole. RMS value of the back electromotive force (EMF). Number of pole pairs. Armature current. Magnetic coercive force. Half slot length in the moving direction. Height where the armature coil is wound. Packing factor of coil. Dimension of the armature coil. Manuscript received January 31, 2013; revised May 1, 2013 and June 25, 2013; accepted July 6, Date of publication July 24, 2013; date of current version February 7, J.-S. Shin, R. Watanabe, and T. Koseki are with the Department of Electrical Engineering, School of Engineering, The University of Tokyo, Tokyo , Japan ( jungseobshin1008@koseki.t.u-tokyo.ac.jp; ryuji@koseki.t.u-tokyo.ac.jp; koseki@koseki.t.u-tokyo.ac.jp). H.-J. Kim is with KOVERY Company, Ltd., Suwon , Korea ( kovery@kovery.com). Color versions of one or more of the figures in this paper are available online at Digital Object Identifier /TIE I. INTRODUCTION LINEAR MOTION is an essential requirement in many applications in industrial fields such as manufacturing processes [1], transportation [2], and robotics [3]. Two methods are generally used to achieve linear motion. One method involves the use of rotary motors with rotaryto-linear converters. Because both rotary and linear motors have been extensively studied, such converters can be designed at low costs and with large thrust densities. However, such mechanical conversion creates large noise and friction, which eventually leads to the deterioration of positioning accuracy. The second method is to employ linear motors. Here, a direct linear drive is possible, which can yield low noise, easy maintenance, and high positioning accuracy. Several types of linear motors are available. In particular, because of the advent of rare-earth permanent magnets [4], a permanent-magnet linear synchronous motor (PMLSM) has been used in applications that require large thrust densities and high positioning accuracies. However, one of the drawbacks of the PMLSM is that the thrust density is comparatively lower than that of rotary motors equipped with rotary-to-linear converters; this drawback has severely limited the high-performance applications of the PMLSM. Hence, increasing the thrust density is one of the most important research topics with regard to the PMLSM today. A transverse-flux-type topology is an ideal alternative in which the flux is carried in the iron back in a plane transverse (perpendicular) to the direction of motion and current flow. Therefore, the magnetic and electric loads in the machine are along different planes. This allows for an inverse relationship to exist between the capability of the machine to produce force and the pole pitch, which results in a higher thrust density [5]. However, the process for manufacturing conventional transverse-flux-type topologies is generally difficult because of the complex structure resulting from the presence of a 3-D magnetic circuit [6]. In conventional manufacturing processes, a large number of segmented components are needed to form a magnetic circuit, and the use of lamination is difficult; this limits its application in industrial fields. In earlier works, a transverse-flux-type cylindrical linear synchronous motor using generic armature cores for rotary machinery has been proposed to address the problem of complex structures in conventional transverse-flux-type topologies [7] IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See for more information.

2 SHIN et al.: LINEAR SYNCHRONOUS MOTOR USING GENERIC ARMATURE CORES FOR ROTARY MACHINERY 4347 the magnetic balance between the armature and field sides is maintained. The fundamental principle of the proposed model is the generation of a longitudinal force that can drive the armature side toward the z-direction. B. Structure and Operational Principle Fig. 1. Fundamental principle of the proposed model. (a) Cross section of a general rotary synchronous motor. (b) Cross section of the proposed linear motor. However, in these works, the structural advantages of the proposed configuration have not been investigated. Furthermore, the design results have not been experimentally validated. This paper mainly focuses on three things. First, the operational principle and structural advantages of the proposed model are investigated. Second, the thrust design and cogging force are evaluated by using an application in which large thrust density and small cogging force are required. Third, the prototype model is experimentally verified. In Section II, the operational principle and structural advantages are introduced. In Section III, we explain a method that can reduce the cogging force of the proposed model. In Sections IV and V, the proposed model is both theoretically and numerically designed by using a magnetic-circuit method and a 3-D finite-element (FE) method (FEM), respectively. Finally, in Section VI, the results and efficacy of the structural concept of our initial prototype model are experimentally validated. II. OPERATIONAL PRINCIPLE AND STRUCTURAL ADVANTAGES A. Fundamental Principle of the Proposed Model The fundamental principle of the proposed model is shown in Fig. 1 [7]. For conventional rotational synchronous machinery, as shown in Fig. 1(a), the number of armature poles is different from that of the field magnets. Therefore, the rotor rotates after the application of an armature current that has a phase difference of 120 with each armature coil. However, if the number of armature poles is equal to that of the field magnets, as shown in Fig. 1(b), the number of magnetic circuits that can be generated is the same as the number of field magnets, and the rotor remains stationary since Fig. 2 shows the configuration of the three-phase unit of the proposed model. In the overall configuration, the armature side is the mover, and the field side is the stator, as shown in Fig. 2(a). The armature side consists of armature units in a nonmagnetic material box. The field side consists of field units inside a stainless steel pipe fixed at both ends. A basic armature unit has an even number of salient poles and a concentrated winding structure along with a four-salient-pole configuration, as shown in Fig. 2. Every armature coil is wound in series with a phase difference of 180. By applying a current to these coils, each of them is excited with a phase difference of 180. In Fig. 2(b), U denotes the current component shifted by 180 from U. A field unit consists of an even number of field magnets (equal to the number of salient poles in the armature unit) and an iron core, as shown in Fig. 2(b). The field magnets are magnetized along the radial direction, similar to that in a conventional cylindrical structure. In this configuration, not only the armature core of three-phase machines but also the armature core of the stepping motor can be used as the armature core. Furthermore, the field side is designed according to the shape of the armature core. When a field unit is accurately located in the center of the armature unit, a magnetic circuit is formed, as shown in Fig. 2(c). The main flux flows transversely from the north pole to the south pole along the armature core. In this manner, the cross-sectional symmetrical magnetic form of the balanced magnetic circuits, which is the same number of salient poles in armature unit, emerges. The armature cores are arranged along the direction of movement (i.e., the z-direction), as shown in Fig. 2(d). Each core is spatially separated by a difference of 120. The field magnets are arranged along the z-direction, and each magnet is electrically separated by 180. The nonmagnetic material spacer isolates the magnetic paths between the iron cores. In this structure, each unit in the armature and field sides is magnetically separated. The flux flow is transverse to the direction of movement, and the longitudinal flux flow (similar to that in a conventional cylindrical structure) back to the common core is absent [8], [9]. Therefore, any core pole combination, including not only the fundamental three-core two-pole combination but also the nine-core eight-pole combination, can be easily achieved by arranging each unit in the armature and field sides along the direction of movement. Fig. 3 shows the principle of generating thrust when a threephase ac current is applied to the armature coil in each phase in the fundamental three-core two-pole combination. The principle of generating thrust in the proposed model is basically the same as that in a conventional PMLSM. When the U-phase core is between the north and south poles where the electric angle of the U-phase core is equal to 90, the armature current

3 4348 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 61, NO. 8, AUGUST 2014 Fig. 3. Principle of generating thrust. (a) Three-phase ac current. (b) Interaction between the armature and field sides. (In (b), only a salient pole in the armature core for each phase is considered because of its symmetrical structure. Furthermore, the iron cores in the field side have been removed for better understanding and clarity.) the V - and W -phases. After interaction with the field magnets, the armature side moves along the z-direction, as shown in Fig. 3(b). In this manner, the three-phase ac current applied to all armature coils in each phase generates the moving magnetic field through all salient poles; thereafter, this moving magnetic field interacts with the field magnets. As a result, a longitudinal force is generated that drives the armature side along the z-direction. Fig. 2. Configuration of the three-phase unit. (a) Entire configuration. (b) Armature and field units. (c) Magnetic circuits. (d) Configuration along the direction of movement. (In (d), the iron cores in the field side have been removed for better understanding and clarity.) with magnitude I is applied to the armature coil in the U-phase, and the armature current with magnitude 0.5I is applied to the armature coils in the V - and W -phase coils in the direction opposite to the U-phase, as shown in Fig. 3(a). Under this condition, the north pole is generated at the salient pole in the U-phase, and the south pole is generated at the salient pole in C. Structural Advantages With regard to the PMLSM research, the structural aspect has to be considered because it considerably affects the performance and stiffness of the linear drive system and is also related to the versatility of its applications in industrial fields. The proposed model has structural advantages because of the following reasons. 1) Simple structure: The magnetic structure comprises only one cylindrical field unit and one unsegmented armature core. Because of the unsegmented armature core, the mechanical stiffness is high. 2) Easy assembly: Since these magnets can be easily placed into the holes of the iron core, there is no need for using a strong adhesive, and specific equipment is not needed to fix these magnets. 3) Use of lamination: Unlike the configuration in conventional transverse-flux-type topologies, the proposed model has a 2-D magnetic circuit in which the main flux

4 SHIN et al.: LINEAR SYNCHRONOUS MOTOR USING GENERIC ARMATURE CORES FOR ROTARY MACHINERY 4349 flows transversely from the north pole to the south pole along the armature core, as shown in Fig. 2(c). In this magnetic circuit, the iron cores in the armature and field units can be easily fabricated using laminated steel plates that are arranged along the z-direction. 4) Cancellation of the strong normal attractive force between the armature and field sides: The strong normal attractive force between the armature and field sides is one of the most important structural problems in conventional single-sided PMLSM; furthermore, it considerably affects the mechanical stiffness of a linear drive system. Therefore, the burden placed on the supporting mechanism is large, thereby complicating the manufacturing process [4]. However, the proposed model is inherently compensated by the cross-sectional symmetrical magnetic form of the balanced magnetic circuits. As a result, mechanical support can be easily achieved. Therefore, the burden placed on the supporting mechanism is small, thereby simplifying the manufacturing process. III. CONSIDERATION OF COGGING FORCE REDUCTION A cogging force arises from the interaction between the field magnet and the armature core. Large cogging forces result in thrust ripples and noise, which result in poor positioning accuracy. In applications such as the driving source in liquid-crystaldisplay color filter inspection systems in which positioning accuracy of less than 5 μm is required, cogging force reduction is one of the most important factors to be considered in the PMLSM design. Many methods have been proposed regarding the reduction of the cogging force in PMLSMs, e.g., skewing, semiclosed slots, and optimization of the magnet length [10], [11]. However, these methods can be a burden during the manufacturing stage and can occasionally affect the manufacturing cost. The magnitude of the cogging force is inversely proportional to the least common multiple (LCM) of the number of cores and poles. Therefore, the cogging force can be significantly reduced by properly selecting a core pole combination. The nine-core eight-pole combination (in which the LCM is four times that of the nine-core six-pole combination based on the fundamental three-core two-pole combination) has been verified as a useful method for cogging force reduction; using this method, the cogging force is reduced to over 50% as compared to the other combinations based on the fundamental three-core two-pole combination [12]. Fig. 4 shows the characteristics of nine-core six-pole and nine-core eight-pole combinations. The former consists of only three sets of fundamental three-core two-pole combinations, as shown in Fig. 4(a). In this configuration, the coil connection is U, V, and W because the V - and W -phase cores in each set have electric phase differences of 120 and 240 with the U-phase core in each set, respectively. In the nine-core eight-pole combination, the position relationship between the armature cores and field magnets is the same as that in the nine-core six-pole combination. However, the sequence of placement of the cores and magnets is different. Fig. 4. Characteristics of two types of core pole combinations. (a) Configuration of nine-core six-pole combination. (b) Configuration of nine-core eightpole combination. (c) Coil connection of the proposed model. (d) Average effective flux in the U-phase. ( U, V,and W are the current components shifted by 180 from their U, V,andW counterparts, respectively. Furthermore, in (a) and (b), only a salient pole in the armature core for each phase is considered because of its symmetrical structure.) As shown in Fig. 4(b), nine armature cores are placed alongside eight magnets for a total distance of T. If the pole pitch is 9 mm, the total distance is 72 mm, and the slot pitch is 8 mm. For this position relationship, the fourth and seventh cores have electric phase differences of 120 and 240 with the first core, respectively. Furthermore, the fifth, sixth, eighth, and ninth cores have electric phase differences of 120 and 240 with the second and third cores, respectively. Therefore, if the first and third

5 4350 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 61, NO. 8, AUGUST 2014 cores are U-phase cores, the fourth and sixth cores are V -phase cores, and the seventh and ninth cores are W -phase cores; this implies that three continuous cores have the same phase. In the proposed model, a Y-connection is used in order to apply a three-phase ac current to the armature coil, as shown in Fig. 4(c). Every armature coil in each core in each phase is wound to all the salient poles in the series with a phase difference of 180. Furthermore, from the position relationship between the armature cores and field magnets in which a south pole is present below the U2 core, the armature coils in U2, V 2, and W 2 have to be wound in the opposite direction to the other cores in the same phase. The armature coil in each core for each phase is connected in parallel. On the other hand, the disadvantage of applying a ninecore eight-pole combination is a decrease in the thrust density. This is a result of the asymmetric characteristics between the armature cores and field magnets, i.e., U1 and U3 cores have an electrical phase difference of 20, as shown in Fig. 4(b). Hence, the average effective flux in the U-phase decreases by 4%, as shown in Fig. 4(d). As a result, the thrust in a nine-core eightpole combination is approximately 4% less than that in a ninecore six-pole combination. However, we have decided to use a nine-core eight-pole combination in the proposed model for cogging force reduction because the decrease in the cogging force is much larger than that in the thrust density. IV. PRELIMINARY DESIGN OF THRUST We have employed the magnetic-circuit method for theoretical modeling of thrust in the preliminary design stage and made the following assumptions [13], [14]. 1) The field unit is accurately located in the center of the armature unit. 2) The permeability of iron cores is infinitely large. Magnetic saturation is neglected. 3) The slot effect is compensated by the Carter coefficient. 4) The flux leakage is neglected. All flux from the field magnet flows into the armature teeth through an air gap. 5) Behavior in a one-phase configuration is considered because behavior in a three-phase configuration can be estimated from the results in the one-phase configuration. A. Air-Gap Flux Density If the slot effect is compensated by the Carter coefficient, the air-gap flux under no-load condition by moving an armature unit φ g (z) is distributed, as shown in Fig. 5. In Fig. 5, z is the distance in the moving direction and can be expressed by velocity v multiplied by time t. Under these conditions, φ g (z) can be expressed by a Fourier series, as shown in φ g (z)= k=1 4φ g (2k 1)π sin ( (2k 1)πa 2τ p ) cos ( (2k 1)πz τ p ). From (1), the fundamental component of φ g (z) and its density B g (z) are expressed in (2) and (3). In (3), B g has been (1) Fig. 5. Air-gap flux distribution by moving an armature unit. derived on the basis of the magnetic-circuit method [13] and is expressed in (4) φ g (z) = 4φ ( ) ( ) g πa πz π sin cos (2) 2τ p τ p B g (z) = φ g(z) = 4B ( ) ( ) g πa πz A g π sin cos (3) 2τ p τ p where B g = ( Ag A m B r + μ rml gc l m ). (4) If the dimensions of the air gap and air-gap length are constant and the flux leakage is neglected, the air-gap flux under no-load conditions is affected by A m and l m. The effective design for large air-gap flux is to select A m and l m as large as possible. This affects the design for large magnetomotive force (MMF) and relatively small magnetic reluctance of the field magnet, which results in a large flux in the magnetic circuit. However, A m and l m cannot be infinitely large, owing to factors such as limited space in the field side, manufacturing cost, and mechanical strength. Therefore, proper selection of A m and l m in the designated volume is important in the actual design stage. B. Maximum Thrust If the flux leakage is negligible, all flux from the field magnets is linked with armature coils, and the back EMF can be expressed as E(z) = N dφ g(z). (5) dt In the control, when the d-axis current is maintained at zero, the maximum thrust per armature unit F t_1unit can be expressed as shown in (6). The maximum thrust is generated in a position in which the electrical phase difference between the armature core and north pole is 90 F t_1unit = p E rmsi = 2 ( ) 2pπφ g NI πa sin. (6) v τ p 2τ p From (6), thrust is generally determined by electrical load NI multiplied by the effective flux flowing to the armature core from the magnetic load H c l m if other conditions are constant. The maximum thrust per three-phase F t_3phase, considering several core pole combinations, is estimated from (7). In (7),

6 SHIN et al.: LINEAR SYNCHRONOUS MOTOR USING GENERIC ARMATURE CORES FOR ROTARY MACHINERY 4351 TABLE I MAIN DESIGN SPECIFICATIONS AND MATERIALS [16], [17] Fig. 6. Three-dimensional mesh FE model. 1.5 specifies the total amount of thrust for a general three-phase machine, and m is the number of armature units per phase. Also, α is a coefficient that denotes the ratio of the average effective flux in the U-phase in core pole combinations which have the asymmetric characteristics between the armature cores and field magnets to the average effective flux in the U-phase in core pole combinations that are based on fundamental threecore two-pole combinations. In the case of a nine-core eightpole combination, α is 0.96, as shown in Fig. 4(d). This is the result of the asymmetric characteristics between the armature cores and field magnets in nine-core eight-pole combination, i.e., U 1 and U 3 cores in nine-core eight-pole combination have an electrical phase difference of 20, as shown in Fig. 4(b), compared with that in nine-core six-pole combination F t_3phase =1.5 F t_1phase =1.5 m α F t_1unit. (7) V. T HRUST DESIGN USING 3-D FEM In spite of our theoretical modeling, all flux from the field magnet may, in reality, not flow into the armature teeth through the air gap. Therefore, a design in the initial design stage employing numerical tools as part of the design process provides useful information regarding flux leakage, flux distribution, and magnetic saturation. In the numerical design and analysis using 3-D FEM, the JMAG-Designer h commercial package is used [15]. The 3-D mesh of the proposed model is shown in Fig. 6. In the 3-D FEM analysis, a four-salient-pole model is considered, and three armature cores per phase and a periodic boundary condition have been applied to save the computation time. A. Specification and Materials in the Proposed Design Table I and Fig. 7 show the primary design specifications and materials used in the proposed model. A total volume of 60 mm 60 mm 108 mm and a ninecore eight-pole combination were selected. In this volume, the Fig. 7. Main design specifications of parts. (a) x y plane. (b) x z plane. pole pitch τ p and slot pitch τ s are 13.5 and 12 mm, respectively. All dimensions in the proposed model, except for d a, are fixed and selected on the basis of spatial limitations and mechanical strength. The core of the 50JN230 in which the maximum magnetic flux density in the linear operation region is closed to 1.5 T and the thickness of laminated piece is 0.5 mm has been used as iron core in the armature and field sides [16]. Also, the permanent magnet of N50M in which B r is about 1.4 T has been used as the field magnet [17]. The velocity v is 1 m/s, and the drive frequency f is Hz.

7 4352 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 61, NO. 8, AUGUST 2014 B. Key Design Variables for Thrust Design When only the structure and total volume are known, it is important to find the point at which the maximum thrust is generated. As shown in (6), thrust is proportional to the multiplication of magnetic and electric loads. Magnetic load H c l m represents the MMF of the field magnet. Electric load NI represents the MMF of the armature side. When the structure and total volume are determined, both loads depend on the space of the total volume. For the design of maximum thrust in the proposed model, we have determined l m to be 2.9 mm on the basis of spatial limitations and mechanical strength in the field unit. The key design variable in thrust design is d a. d a denotes the half slot length in the moving direction, and an armature coil is wound in d a. It is related to the electric load, as expressed as (8). In (8), N(d a,l c ) is the number of windings per armature pole, and A w (d a,l c ) is the cross-sectional area where the armature coil is wound. Geometrically, the larger d a is, the higher the winding turn per armature pole can be achieved. This represents the increase of electrical load by increasing MMF N(d a,l c )I = k cpa w (d a,l c )I (8) S c where A w (d a,l c )=d a l c. (9) However, the term d a affects the width l z and cross section A a of the armature core, as expressed in (10). In (10), the larger the value of d a under the same τ s, the smaller A a (d a,l t ) is, which means a decrease in the amount of effective flux flowing to the armature core. Therefore, the design for maximum thrust in the designated volume should find the point where the multiplication of spatially determined effective flux and electric loads reaches the maximum value [18] A a (d a,l t )=l z (d a ) l t (10) where l z (d a )=τ s 2d a. (11) C. Total Longitudinal Force Total longitudinal force per phase acting in moving direction consists of cogging force and thrust [7]. Here, let us define that the maximum thrust per phase at a certain design point F t_1phase is obtained from the difference between the total longitudinal force F total_1phase and the cogging force F c_1phase in the mover position where the maximum thrust is generated, as expressed in (12). Cogging force has been calculated by applying the virtual displacement principle, as expressed in (13). In (13), W is the magnetic energy stored in a phase, which is calculated as the consequence of the numerical electromagnetic field calculation F t_1phase = T total_1phase F c_1phase (12) F c_1phase = dw dz (13) φ=const. Fig. 8 and Table II show the 3-D FEM results of thrust per phase under the rated condition where I =4Aand the winding Fig. 8. Fig. 9. Three-dimensional FEM results of thrust. TABLE II WINDING TURN PER ARMATURE POLE Ratio of cogging force to thrust. turn per armature pole. The maximum thrust is at d a =3.5 mm and approximately 27 N. Thrust decreases from d a =3.5 mm, which means decreasing flux linkage to the armature coil, regardless of spatially increased MMF on the armature side. However, in selecting the design of a PMLSM, cogging force is also an important aspect because large cogging force causes thrust ripple, which results in poor positioning accuracy. Thus, it is important to find the point at which large thrust and low cogging force can be achieved. We have evaluated this aspect using the ratio of cogging force to thrust. Fig. 9 shows the ratio of cogging force to thrust based on 3-D FEM values. From the results, d a =2.5 mm is a good value when considering a design for large thrust and low cogging force within a designated dimension. For that reason, we have decided that d a =2.5 mm is the best point in the initial design. VI. FUNDAMENTAL EXPERIMENT FOR THE VALIDATION OF RESULTS IN THE DESIGN POINT On the basis of the design point in the initial design stage, we have fabricated the initial prototype model. In this section, the efficacy of our structural concept is experimentally validated. In the experiment, resistance, inductance, back EMF, thrust, and cogging force are measured and analyzed.

8 SHIN et al.: LINEAR SYNCHRONOUS MOTOR USING GENERIC ARMATURE CORES FOR ROTARY MACHINERY 4353 TABLE III RESISTANCE AND INDUCTANCE Fig. 12. Measuring equipment for thrust. Fig. 10. Prototype model and measuring equipment for back EMF. Fig. 11. Three-dimensional FEM and empirical results of back EMF. A. Resistance and Inductance Table III shows the measurement results of resistance and inductance. They were measured using a digital multifunctional meter and LRC meter at room temperature (22.7 C). B. Back EMF The prototype model and the equipment used to measure back EMF are shown in Fig. 10. The back EMF is measured from the open-circuit voltage. The three-phase voltage waveforms were recorded using a digital oscilloscope (TECTONICS-TDS 3034C). Fig. 11 shows the 3-D FEM and empirical back EMF with the mover position at v =1m/s. The 3-D FEM data were calculated from flux linking with the armature core at no-load conditions. There is relatively good agreement between 3-D FEM and empirical values, and both waveforms are close to sinusoidal. C. Thrust The equipment used to measure thrust basically consists of a load cell, a potential meter, a dc amplifier, and a data collector, Fig. 13. Three-dimensional FEM and empirical results of thrust. (a) Thrust at I =4A. (b) Thrust armature-current characteristics. (In (b), the data in parentheses in the x-axis denote the current density.) as shown in Fig. 12. The placement of the prototype was carefully planned to prevent measuring error. Thrust was measured using a load cell at different mover positions when dc current was applied to the armature coil. The displacement of mover was measured using a potential meter, and the data from the load cell and potential meter were recorded in the data collector. Fig. 13 shows the thrust under the rated condition where I =4A and thrust armature-current characteristics. There is a good agreement between 3-D FEM and empirical values. Empirical values are approximately equal to 3-D FEM values. These results prove the validity of the design point that was obtained from the design developed using 3-D FEM.

9 4354 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 61, NO. 8, AUGUST 2014 TABLE IV COMPARISON OF THRUST DENSITY Fig. 14. Three-dimensional FEM and empirical results of cogging force. D. Cogging Force In this research, the method used to measure cogging force was basically the same as that used for thrust, except for the armature current. Cogging force was measured by the load cell at different mover positions under no-load conditions. The displacement of the mover was also measured using the potential meter, and the data from load cell and potential meter were recorded in the data collector. Fig. 14 shows the 3-D FEM and empirical results of cogging force. Large cogging force per phase is significantly reduced by applying the nine-core eight-pole combination. The maximum value of total cogging force was approximately 5 N, which was approximately 6.4% of the thrust in the rated region. However, considering the performance requirement for applications in which a positioning accuracy of less than 5 μm is required, further improvement of cogging force remains to be desired. E. Thrust Density Thrust density is an important aspect to evaluate. We have calculated thrust density on the basis of the following to compare it with other proposals for commercial products [19], [20]. To compare thrust density, rated thrust has been used. Table IV shows the thrust density in the proposed model and four other commercial products. 1) F volume : thrust per total volume V t in which the armature side is faced with the field side. 2) F dimension : thrust per total dimension S t of active air-gap area. 3) F weight : thrust per total mover weight W t. It is difficult to evaluate the superiority or inferiority of not only the proposed model but also other proposed models on account of insufficient data about the measuring methods, drive conditions, current density, cooling methods, etc. However, results from the initial prototype model, which is based on the proposed configuration, are not in the top tier even though they are in the range of the latest linear motor performance. The small thrust density results from low slot fill factor in the initial prototype model. The number of winding turns in a salient pole in the proposed model was determined by the slot in the moving direction and space around a salient pole in the cross section in the x y plane, as shown in Fig. 2. The slot in the moving direction was determined by slot pitch, and the space around a salient pole in the cross section in the x y plane was determined by the number of salient poles in the armature core. Compared with the slot fill factor in a slot with a moving direction of 0.75 in the initial prototype model, the slot fill factor in the cross section of the x y plane is approximately 0.16, which indicates that there is a large amount of wasted space. Therefore, for large thrust density, the optimal ratio between pole pitch and the number of salient poles in the armature core should be considered. From this result, our future work, which will focus on structural and design improvements for large thrust density, has been clarified. VII. CONCLUSION In this paper, a transverse-flux-type cylindrical linear synchronous motor using generic armature cores for rotary machinery to address the problem of the complex structure in transverse-flux-type topologies has been proposed. The advantages of the proposed configuration are the following: 1) simple structure; 2) easy assembly; 3) the use of lamination; 4) cancellation of the strong normal attractive force between the armature and field sides. These advantages are helpful to facilitate the manufacturing of PMLSMs with transverse-flux-type topology. In the initial design stage, the design for thrust and cogging force in an application in which large thrust and small cogging force are required was undertaken using a magneticcircuit method and 3-D FEM. Thrust is proportional to the

10 SHIN et al.: LINEAR SYNCHRONOUS MOTOR USING GENERIC ARMATURE CORES FOR ROTARY MACHINERY 4355 multiplication of electric and magnetic loads; therefore, a design that optimizes maximum thrust in the designated volume requires finding the point at which the multiplication of spatially determined effective flux and electric loads reaches the maximum value. However, in selecting the design of a PMLSM, cogging force should also be considered because large cogging force causes thrust ripple, which results in a poor positioning accuracy. Therefore, the point at which large thrust and small cogging force could be achieved was selected as the critical design point in the initial design stage. The initial prototype model was fabricated to validate the results for the selected design point and our structural concept, and a fundamental experiment was carefully conducted. A good agreement between 3-D FEM and empirical values proved the validity of the design point and the effectiveness of our structural concept. In addition, it was verified that a nine-core eightpole combination was useful for significant reduction in the proposed model. Although it was verified that the results from the initial prototype model were close to the latest linear motor technology, improvements of thrust and cogging force remain to be achieved. Therefore, in the future, we will propose structural and design improvements for high performance, including large thrust density and small cogging force. REFERENCES [1] H. W. Chow and N. C. Cheung, Disturbance and response time improvement of submicrometer precision linear motion system by using modified disturbance compensator and internal model reference control, IEEE Trans. Ind. Electron., vol. 60, no. 1, pp , Jan [2] K. Suzuki, Y. J. Kim, and H. 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Magn., vol. 48, no. 11, pp , Nov [19] Yasukawa Electric Corp., [Online]. Available: co.jp/ [20] GMC Hillstone Co. Ltd., [Online]. Available: product/shaft.html Jung-Seob Shin (M 11) received the M.S. degree in electrical engineering from The University of Tokyo, Tokyo, Japan, in 2011, in the laboratory of Prof. T. Koseki, where he is currently working toward the Ph.D. degree in electrical engineering. His interests include linear synchronous motors, electrical machinery design, and motor control. Mr. Shin is a member of the IEEE Industrial Electronics Society. Ryuji Watanabe (S 13) received the B.S. degree in electrical engineering from The University of Tokyo, Tokyo, Japan, in 2012, where he is currently working toward the M.S. degree in electrical engineering. His research interests include linear synchronous motors, synchronous generators, and electrical machinery design. Takafumi Koseki (S 87 M 92) received the Ph.D. degree in electrical engineering from The University of Tokyo, Tokyo, Japan, in He is currently an Associate Professor with the Department of Electrical Engineering, School of Engineering, The University of Tokyo. His current research interests include public transport systems, particularly linear drives, and the analysis and control of traction systems. Prof. Koseki is a member of the Institute of Electrical Engineers of Japan, Japan Society of Mechanical Engineering, Japan Society of Applied Electromagnetics and Mechanics, Japan Society of Precision Engineering, and Japan Railway Electrical Engineering Association. Houng-Joong Kim received the Ph.D. degree in electrical engineering from Musashi Institute of Technology, (currently Tokyo City University), Tokyo, Japan, in He was a Researcher with the Hitachi Research Laboratory, Hitachi Ltd., Hitachi, Japan, until 2009, where he was involved in linear drive research and invented the tunnel actuator (TA). This design is used to create the globally fastest drilling machine for printed circuit boards. He is currently the CEO of KOVERY Company, Ltd., Suwon, Korea. His current research interests include linear motor designs and precise drive controls for industrial applications. Dr. Kim is a member of the Korean Institute of Electrical Engineers. He was a recipient of an R&D 100 Award for the TA in 2007.