Linear Quadratic Optimal Control Topics
|
|
- Blaze Floyd
- 6 years ago
- Views:
Transcription
1 Linear Quadratic Optimal Control Topics Finite time LQR problem for time varying systems Open loop solution via Lagrange multiplier Closed loop solution Dynamic programming (DP) principle Cost-to-go function computed from DP Infinite time LQ problem for LTI systems Convergence of P(t, t f ) Closed loop stability P as solution of ARE via Hamiltonian matrix Selection of Q, R and S Robustness (Return difference inequality) 1 Symmetric root-locus and cheap control 2 Some extensions Discrete time LQ Pole-placement within a pre-defined region Frequency shaping 1 Not covered in Spring Not covered in Spring 2008 M.E. University of Minnesota 212
2 Linear Quadratic (LQR) Optimal Control - Motivation Pole-placement approach allows ones to choose where to place the poles SI feedback gain unique MI feedback gain non-unique (e.g. need Hautus- Keyman Lemma or eigenvector placement) Main issue: where should we place the poles??? Should consider trade-off between performance, robustness and control effort. LQ technique tries to do some trade-off without specifying desired poles locations M.E. University of Minnesota 213
3 Finite Time Linear Quadratic Optimal Regulator Problem m input, u R m, n state system with x R n : ẋ = A(t)x + B(t)u; x(0) = x 0. (27) Find open loop control u(τ), τ [t 0, t f ] such that the following objective function is minimized: J(u,x 0, t 0 ) = 1 2 xt (t f )Sx(t f )+ 1 2 tf t 0 [ x T (t)q(t)x(t) + u T (t)r(t)u(t) ] dt (28) Q(t) = Q T (t) and S are symmetric positive semidefinite n n matrices R(t) = R T (t) is a symmetric positive definite m m matrix. Notice that x 0, t 0, and t f are fixed and given data. M.E. University of Minnesota 214
4 The control goal generally is to keep x(t) close to 0 3, especially, at the final time t f, using little control effort u. To wit, notice in (28) x T (t)q(t)x(t) penalizes the transient state deviation, x(t f ) T Sx(t f ) penalizes the finite state u T (t)r(t)u(t) penalizes control effort. Output regulation: If y = C(t)x is the output, we can define: Q(t) = C T (t)w(t)c(t) where W(t) is a symmetric, positive definite output weighting matrix. 3 LQ can be modified for the trajectory tracking case M.E. University of Minnesota 215
5 General Finite Time Optimal Control Plant: ẋ = f(x,u, t); x(t 0 ) = x 0 given. Time interval: t [t 0, t f ]. Cost function to be minimized: J(u( ), x 0 ) = φ(x(t f )) + tf t 0 L(x(t), u(t), t)dt First term is the final cost and the second term is the running cost. Problem: Find u(t), t [t 0, t f ] such that J(x 0, u( )) is minimized, subject x(t) satisfying the plant equation x(t 0 ) = x 0 given. Solution is to convert constrained optimal control into unconstrained optimal control using Lagrange multiplier λ(t) R n : J(u, x 0 ) = J(u( ), x 0 ) + tf t 0 λ T (t)[f(x,u,t) ẋ]dt. M.E. University of Minnesota 216
6 Note that d dt (λt (t)ẋ(t)) = λ T (t)x(t) + λ T (t)ẋ. So tf t 0 λ T ẋdt = λ T (t f )ẋ(t f ) λ T (t 0 )ẋ(t 0 ) tf t 0 λt xdt. Let us define the so called Hamiltonian function H(x, u, t) := L(x, u,t) + λ T (t)f(x,u,t). Necessary condition for optimality: Variation of the modified cost δ J with respect to all feasible variations δx(t) and δu(t) and δλ(t) should vanish. Using integration by parts: λẋ = λx λx, J = φ(x(t f )) λ T (t f )x(t f ) + λ T (t 0 )x(t 0 ) + tf t 0 H(x(t), u(t), t) + λ(t)x(t)] dt M.E. University of Minnesota 217
7 δ J = [φ x λ T ]δx(t f ) + λ T (t 0 )δx(t 0 ) + + tf t 0 [H x + λ T ]δx + H u δudt tf t 0 δλ T [f(x(t), u(t), t) ẋ]dt Since x(t 0 ) = x 0 is fixed, δx(t 0 ) = 0. Otherwise, other variations δx(t), δu(t) or δλ(t) are all feasible. Hence, λ = H x = L f λt x x (29) ẋ = f(x, u,t) (30) H u = L u λt f u = 0 (31) λ T (t f ) = φ x (x(t f)) (32) x(t 0 ) = x 0. (33) This is a set of 2n differential equations (in x and λ) with split boundary conditions at t 0 and t f : x(t 0 ) = x 0 and λ T (t f ) = φ x (x(t f )). M.E. University of Minnesota 218
8 Finite Time LQ Regulator Solution Open loop formulation: With L(x, u, t) = x T (t)q(t)x(t) + u T (t)r(t)u(t) φ(x(t f )) = x T (t f )Sx(t f ) f(x, u, t) = A(t)x + B(t)u Let λ(t) R n be the Lagrange multiplier. Using the above definitions in Eqs.(29)-(33), the optimal control is given by (see (31)): u o (t) = R 1 B T (t)λ(t) where λ(t) and x(t) satisfy the Hamilton-Jacobi equation ((29)-(30)): (ẋ ) ( )( A(t) B(t)R = 1 B T (t) x λ Q(t) A T (t) λ) }{{} Hamiltonian Matrix - H(t) with boundary conditions given by (see (32)-(33)): x(t 0 ) = x 0 ; λ(t f ) = Sx(t f ). (34) M.E. University of Minnesota 219
9 Boundary conditions specified at initial time t 0 and final time t f (two point boundary value problem). In general, these are difficult to solve requires iterative methods such as shooting method. Optimal control is open loop. It is computed by first computing λ(t) for all t [t 0, t f ] and then applying u o (t) = R 1 B T (t)λ(t). Open loop control is not robust to disturbances or uncertainties. M.E. University of Minnesota 220
10 Closed loop control solution Consider X 1 (t) R n n and X 2 (t) R n n satisfying the Hamilton-Jacobi equation: ) ( ) (Ẋ1 A(t) B(t)R = 1 B T (t) Ẋ 2 Q(T) A T (t) }{{} Hamiltonian Matrix - H ( X1 X 2 ) with X 1 (t f ) non-singular (e.g. X 1 (t f ) = I n n ), and X 2 (t f ) = SX 1 (t f ). This requires solving the 2n n differential equations backward in time. Claim: Assuming that X 1 (t) is invertible for all t [t 0, t f ]. Then, we can express x(t) and λ(t) satisfying the Hamilton-Jacobi equation by: ( ) x(t) λ(t) = ( ) X1 (t) v X 2 (t) for some constant v R n. Moreover, λ(t) = [X 2 (t)x 1 1 (t)]x(t) M.E. University of Minnesota 221
11 This can be shown by direct substitution, with v = X1 1 (t 0)x 0, that x(t) and λ(t) satisfy the Hamilton- Jacobi equation, as well as the boundary conditions. This result implies that the optimal control can be expressed as closed loop state-feedback u o (t) = R 1 B T (t)λ(t) = R 1 B T (t)p(t)x(t) where P(t) := X 2 (t)x 1 1 (t) Rn n. Differentiating P(t) := X 2 (t)x1 1 (t) and using Hamilton-Jacobi equation (for X 1 (t) and X 2 (t)), we find that P(t) satisfies the continuous time Riccati differential equation (CTRDE): P(t) = A T (t)p(t) P(t)A(t) + P(t)B(t)R 1 (t)b T (t)p(t) Q(t); (35) with boundary condition P(t f ) = S. P(t) is symmetric and positive semi-definite. It is symmetric because S is symmetric. To show that P(t) is positive semi-definite, we first interpret P(t) in M.E. University of Minnesota 222
12 terms of the performance index. The claim is that the minimum cost is: J(u o, x 0, t 0 ) = 1 2 xt 0 P(t 0 )x 0. where u o (t) is the optimal control. From the optimal control, u o (t) = R 1 (t)b T (t)p(t)x(t), and the form of the quadratic cost function Eq.(28), we know 4 that J o (x 0, t 0 ) = J(u o, x 0, t 0 ) = 1 2 xt 0 P(t 0 )x 0. for some positive semi-definite matrix P(t 0 ). To show that P(t0 ) = P(t 0 ), we need to understand the Dynamic Programming Principle. 4 Note that the closed loop system is linear so that x(t) = Φ(t, t0 )x 0 M.E. University of Minnesota 223
13 Dynamic Programming (DP) Principle Consider a shortest path problem in which we need to traverse a network from state i 0 and to reach state 5 with minimal cost. Cost to traverse an arc from i j is a ij > 0. Cost to stay is a ii = 0 for all i. Since there are only 4 non-destination states, state 5 can be reached in at most N = 4 steps. Total cost is sum of the cost incurred, i.e. if the (non-optimal) control policy π is , then J(π) = a 22 + a 23 + a 34 + a 45 Goal is to find the policy that minimizes J. M.E. University of Minnesota 224
14 As an optimization problem the space of 4 step policy has a cardinality of 5 4 = 625 DP algorithm: We start from the end stage (N = 4), i.e. you need to reach the state 5 in one step. Suppose that you are in state i, the cost to reach state 5 is min{a i5 } = a i5 The optimal and only choice for the next state, if currently at state i, is u (i,n) = 5. The optimal cost-to-go is J (i,n) = a i5. Node u J (i,n) Consider the N 1st stage, and you are in state i. We can have the policy π : i j 5. Since the minimum cost to reach state 5 from state j is J (j, N), the optimal control policy is: min j (a ij + J (j, N)) = min{a i1 + a 15, a i2 + a 25,..., a i5 + a 55 } M.E. University of Minnesota 225
15 For i = 4 (for instance), j a 4j a 4j + J (j, N) Thus, the j the optimizes this is: j = 4 (stay put) so that u (4, N 1) = 4 and J (4, N 1) = 3. Doing this for each i, we have at stage N 1, Optimal policy: u (i,n 1) = arg min j (a ij + J (j, N)) Optimal cost-to-go: J (i,n 1) = min j (a ij + J (j, N)) Node u (i,n 1) J (i,n 1) If we are at the N 2nd stage, and you are in state i, M.E. University of Minnesota 226
16 Optimal policy: u (i,n 2) = arg min j (a ij + J (j, N 1)) Optimal cost-to-go: J (i,n 2) = min j (a ij + J (j, N 1)) Node u (i,n 2) J (i,n 2) Notice that from state 2, the 3 step policy has a lower cost of 4.5 than the 2 step policy with a cost of 5.5. Repeating the propagation procedure for the optimal policy and optimal cost-toabove until N = 1. Then the optimal policy is u (i,1) and the minimum cost is J (i,1). M.E. University of Minnesota 227
17 The optimal sequence starting at i 0 is: i 0 u (i 0, 1) u (u (i 0,1),2) u (u (u (i 0,1), 2),3) 5 Remarks: At each stage k, the optimal policy u (i,k) is a state feedback policy. i.e. it determines what to do depending on the state that you are in. Policy and optimal cost-to-go are computed backwards in time (stage) At each stage, the optimization is done on the space of intermediate states, which has a cardinality of 5. The large optimization problem with cardinality of 5 4 has been reduced to 4 simpler optimization problem with cardinality of 5 each. The tail end of the optimal sequence is optimal - this is the Dynamic Programming Principle. i.e. if M.E. University of Minnesota 228
18 the optimal 4 step sequence π 4 starting at i 0 is: i 0 u (i 0, 1) u (u (i 0,1),2) u (u (u (i 0,1), 2),3) 5 then the sub-sequence π 2 u (i 0, 1) u (u (i 0,1),2) u (u (u (i 0,1),2),3) 5 is the optimal 3 step sequence starting at u (i 0, 1). This is so because if π 3 is another 3 step sequence starting at u (i 0,1) with a strictly lower cost than π 3, then the 4-step sequence i 0 π 3 will also have a lower cost than π 4 = i 0 π 3 which is assumed to be optimal. M.E. University of Minnesota 229
19 Dynamic Programming (DP) Principle Continuous time System: ẋ = f(x(t), u(t), t), x(t 0 ) = x 0, Cost index: J(u( ), t 0 ) = tf t 0 L(x(t), u(t), t)dt + φ(x(t f )). (36) Suppose that u o (t), t [t 0, t f ] minimizes (36) subject to x(t 0 ) = x 0 and x o (t) is the associated state trajectory. Let the minimum cost achieved using u o (t) be: J o (x 0, t 0 ) = argmin u(τ),τ [t0,t f ]J(u( ), t 0 ) Then, for any t s.t. t 0 t + t t f, the restriction of the control u o (τ) to τ [t + t, t f ] minimizes J(u( ), t 0 + t) = tf t 0 + t L(x(t), u(t), t)dt+φ(x(t f )). subject to initial condition x(t 0 + t) = x o (t 0 + t). i.e. u o (τ) is optimal over the sub-interval. M.E. University of Minnesota 230
20 Typical application of DP Solve the optimal control problem for sub-interval [t 1, t f ] with arbitrary initial states, x(t 1 ) = x 1. Let the control that is optimal be u(t) = u o (t, t 1, x 1 ) and let J o (x 1, t 1 ) be the optimal cost given initial state x(t 1 ) = x 1. Now consider t 0 < t 1. The optimal control u o (t, t 0, x 0 ) for the interval [t 0, t f ] with initial states, x(t 0 ) = x 0 is given as follows. For t 0 t t 1, u o (t, t 0, x 0 ) is the u(t) that minimizes: t1 t 0 L(x(t), u(t), t)dt + J o (x(t 1 ), t 1 ) subject to ẋ(t) = f(x(t), u(t), t). Notice that x(t 1 ) is unknown a-priori since it depends on u(t). For t 1 t t f, the optimal control u o (t, t 0, x 0 ) = u o (t, t 1, x(t 1 )) M.E. University of Minnesota 231
21 where x(t 1 ) is the state achieved at t = t 1 from the initial state x 0 using optimal control u o (t, t 0, x 0 ) over the interval [t 0, t 1 ]. This procedure can be repeated by taking the initially time further and further back. Note: The optimal cost J o (x,t) is the cost-to-go function at time t. M.E. University of Minnesota 232
22 Relating P(t) to cost-to-go Let us apply DP to the LQ case (note: without the 1/2 for simplicity): L(x, u, t) = x T Q(t)x + u T R(t)u f(x, u, t) = A(t)x + Bu J = tf t 0 L(x, u,t)dt + φ(x(t f )). At t = t f, the cost-to-go function is simply: Hence, P(tf ) = S. J o (x,t f ) = x T Sx = x T P(tf )x Let t 1 = t f and consider t = t 1 t where t is infinitesimally small. The optimal control at t given the state x(t) is minimize min u(t) L(x, u, t) t + J o (x(t 1 ), t 1 ) Now, x(t 1 ) = x(t) + f(x(t), u(t), t) t. Thus, we M.E. University of Minnesota 233
23 minimize w.r.t. u(t), [ x(t) T Q(t)x(t) + u T (t)r(t)u(t) ] t+ J o (x(t) + [A(t)x(t) + B(t)u(t)] t, t 1 ) [ x(t) T Q(t)x(t) + u T (t)r(t)u(t) ] t + x(t) P(t 1 )x(t) + [x T (t)a T (t) + u T (t)b T (t)] P(t 1 )x(t) t + x T (t) P(t 1 )[A(t)x(t) + B(t)u(t)] t Differentiating w.r.t. u(t), we get the optimal control policy: u ot R(t) + x T (t) P(t 1 )B(t) = 0 u o (t) = R 1 (t)b T (t) P(t 1 )x(t) The updated optimal cost-to-go function is: J o (x(t), t) [ x(t) T Q(t)x(t) + u ot (t)r(t)u o (t) ] t + x(t) P(t 1 )x(t) + [x T (t)a T (t) + u ot (t)b T (t)] P(t 1 )x(t) t + x T (t) P(t 1 )[A(t)x(t) + B(t)u o (t)] t M.E. University of Minnesota 234
24 This shows that J o (x(t), t) x T (t) P(t 1 )x(t) + x T (t) [ A T (t) P(t 1 ) + P(t 1 )A(t) P(t 1 )B(t)R 1 (t)b T (t) P(t 1 ) + Q(t) ] x(t) t = x T (t) P(t)x(t) where ( P(t1 ) P(t) ) = [ A T (t) P(t 1 ) + P(t 1 )A(t) P(t 1 )B(t)R 1 (t)b T (t) P(t 1 ) + Q(t) ] t (37) Thus, we have shown that at t, J o (x(t), t) = x T (t) P(t)x. Let t t 1, t t t and repeat the process and we get the update recursion in Eq.(37). M.E. University of Minnesota 235
25 As t 0, we have Eq.(37) becomes: P(t) = A T (t) P(t) + P(t)A(t) P(t)B(t)R 1 (t)b T (t) P(t) + Q(t); which is exactly the Riccati differential equation as before. Hence P(t) = P(t). Note: Since x T (t)p(t)x(t) = tf t 0 [ x T (τ)q(τ)x(τ) + u T (τ)r(τ)u(τ) ] dτ + x T (t f )Sx(t f ) for any x(t), P(t) is positive semi-definite for any t t f. M.E. University of Minnesota 236
26 Finite time LQ Summary The finite time LQ regulator problem is solved by the control: u (t) = R 1 (t)b T (t)p(t)x(t) (38) where P(t) R n n is the solution to the continuous time Riccati Differential Equation (CTRDE): P(t) = A T (t)p(t) P(t)A(t) + P(t)B(t)R 1 (t)b T (t)p(t) Q(t); with boundary condition P(t f ) = S. P(t) is positive-semi definite The minimum cost achieved using the above control: J (x 0, t 0 ) := min u( ) J(u, x 0 ) = 1 2 xt 0 P(t 0 )x 0 M.E. University of Minnesota 237
27 Remarks 1. The control formulation works for time varying systems, e.g. nonlinear systems linearized about a trajectory. 2. The optimal control law is in the form of a time varying linear state feedback with feedback gain K(t) := R 1 (t)b T (t)p(t), although the control problem is formulated to ask for an open loop control. The open loop optimal control can be obtained, if so desired, by integrating (27) with the control (38). It is, however, much better to utilize feedback than to use openloop. 3. P(t) is solved backwards in time from t f t 0 and should be stored in memory before use. 4. The matrix function P(t) is associated with the socalled cost-to-go function. If at time t, t 0 t t f and the state happens to be x(t), then, the control policy (38) for the remaining time period [t, t f ] is also optimal for the problem (28) J(u,x(t), t, t f ) (i.e. with t 0 substituted by t and x 0 substituted by M.E. University of Minnesota 238
28 x(t)). In this case, the minimum cost is min u J(u,x(t), t) = 1 2 xt (t)p(t)x(t) M.E. University of Minnesota 239
29 Infinite Horizon LQ ẋ = Ax + Bu; x(t 0 ) = x 0 = min J(u,x 0 ) u tf t 0 [x T Qx + u T Ru] dt + x T (t f )Sx(t f ) Solve P(t, t f ) in (35) backwards in time. Does P(t, t f ) exist (i.e. does it converge when t )? If lim t P(t, t f ) = lim tf P(t, t f ) = P does exist, there is a constant state feedback gain given by: K = R 1 B T P. Will the closed loop system: ẋ = (A BK)x be stable? If lim t P(t, t f ) = lim tf P(t, t f ) = P does exist, we know that it must satisfy P(t) = 0, i.e. A T P + P A P BR 1 B T P + Q = 0. (39) M.E. University of Minnesota 240
30 which is called the Algebraic Riccati equation (ARE). In that case, which solution of ARE does the asymptotic solution of (35) correspond to? M.E. University of Minnesota 241
31 Boundedness of P(t, t f ) Solve P(t) backwards in time t = t f according to the CTRDE P(t) = A T (t)p(t) P(t)A(t) + P(t)B(t)R 1 (t)b T (t)p(t) Q(t); with boundary condition P(t f ) = S. In this section, we assume that S = 0. This is reasonable for t 0 since the running cost over an infinite horizon should dominate over cost at the terminal time. See textbook for cases when S 0. Proposition If (A, B) is controllable (or just stabilizable) then for any t < t f, P(t, t f ) < M where M is a positive definite matrix, in that: for all x R n, x T P(t, t f )x < x T Mx Moreover, as T, P(t, t+t) (which is the same as P(t T, t) converges to some matrix positive definite matrix P. Proof: We give the proof for the (A, B) controllable. Let t > 0 be an arbitrary fixed time interval. For M.E. University of Minnesota 242
32 any initial time t < t f t and initial state x 0, we can design a control u(τ), τ [t, t + t] such that x(t + t) = 0; and u(τ) = 0 for τ > t + t. The cost associated with this control is finite and is independent of t. By choosing different x 0, we can define a positive definite matrix M such that x T Mx is the cost for initial state x using the control thus constructed. Secondly, notice that for any > 0, = J(u,t, t f ) = J(u,t, t f + ) tf t [ x T Qx + u T Ru ] tf + [ dτ + x T Qx + u T Ru ] dτ t f } {{ } 0 The optimal cost for the interval [t 0, t f + ] must be greater than or equal to the optimal for J(u, t, t f ), i.e. the optimal cost increases as the time interval increases for the same initial condition. Suppose not, the [t, t f ] portion of the optimal cost for the [t, t f + ] would be less than the supposed optimal for the interval [t, t f ], which is a contradiction. M.E. University of Minnesota 243
33 This shows that for any and x R n, x T P(t, t f )x x T P(t, t f + )x = x T P(t, t f )x x T Mx. From analysis, we know that a non-decreasing, upper bounded function converges, thus, x T P(t, t f + )x converges as. By choosing various x, a matrix P can be constructed s.t. for any x, x T P(t, t f + )x x T P x. M.E. University of Minnesota 244
34 Stability PropositionLet Q = C T C and suppose that (A, B) is stabilizable. If (A, C) is observable (or detectable), then optimal closed loop control system is stable. ẋ = (A BR 1 B T P )x Furthermore, if (A, C) is observable, then P is positive definite. Otherwise, P is positive semidefinite. Proof Suppose that (A, C) is detectable but the closed loop system is unstable. Let ν be the unstable eigenvector of A BR 1 B T P such that λν = (A BR 1 B T P )ν; Re(λ) > 0. Let x(t 0 ) = ν be the initial state. Then, x(t) = e λ(t t 0) ν. Since (A, B) is stabilizable, t 0 x T Qx dt < ; t 0 u T Ru dt < M.E. University of Minnesota 245
35 We assume λ is real below for simplicity. If λ is complex, we need to consider both λ and λ simultaneously. Then, since e λ(t t 0) > 1 for all t t 0 > 0, t 0 ν T Qν e 2λ(t t 0) dt < Cν = 0. t 0 u T Ru dt = ν T [PBR 1 B T P]ν R 1 B T P ν = 0. t 0 e 2λ(t t 0) dt < This implies that (A BR 1 B T P )ν = Aν = λν This contradicts the assumption that (A, C) is detectable, since, ( λi A C ) ν = ( 0 0). Hence, (A, C) detectable, implies that the closed loop system is stable. M.E. University of Minnesota 246
36 To show that P is strictly positive definite when (A, C) is observable, suppose that P is merely positive semi-definite so that, x T 0 P x 0 = t 0 x T C T Cx + u T Ru dt = 0 for some initial state x(t 0 ) = x 0. This implies that for all t, u T (t)ru(t) = 0 or u(t) = 0.; and Cx(t) = 0. Or, for all t, ẋ = Ax; Cx = 0. This is not possible if (A, C) is observable. If (A, C) is merely detectable. Let ν be an unobservable eigenvector. Then, for x(t 0 ) = ν, u = 0 is the optimal control and x(t) = e λ(t t0) ν is the state trajectory, since t o x T (t)c T Cx(t) dt = 0. Thus, ν T P ν = 0. M.E. University of Minnesota 247
37 Solving ARE via the Hamiltonian Matrix For the infinite time horizon LQ problem, with (A, B) stabilizable and (A, C) detectable, the steady state solution of the CTRDE P must satisfy the Algebraic Riccati Equation (ARE) (i.e. by setting P = 0): A T P + P A P BR 1 B T P + Q = 0. (40) This is a nonlinear algebraic quadratic matrix equation. There are generally multiple solutions. Is it possible to solve this without integrating the CTRDE (35)? Recall that the solution P(t) can be obtained from the matrix Hamilton equation: ) ( ) (Ẋ1 A BR = 1 B T Ẋ 2 Q A }{{ T } Hamiltonian Matrix - H ( X1 X 2 ) (41) with boundary conditions: X 1 (t f ) invertible and X 2 (t f ) = SX 1 (t f ) so that P(t) = X 2 (t)x1 1 (t). Denote the 2n eigenvalues and 2n eigenvectors of H by respectively: {λ 1, λ 2,..., λ 2n }, {e 1, e 2,..., e 2n } M.E. University of Minnesota 248
38 Let us choose n pairs of these: ( F Λ = diag {λ i1, λ i2,..., λ in }, G) = ( e i1 e i2... e in ), Proposition Let P := GF 1 where the columns of ( F G) R 2n n are n of the eigenvectors of H. Then, P satisfies the Algebraic Riccati Equation (40). Proof: We know that P(t) = X 2 (t)x1 1 (t) where X 1 (t) and X 2 (t) satisfy the Hamiltonian differential equation (41). For P = GF 1 to satisfy Eq.(40), one needs only show that P(t) = 0 when X 1 (t) = F and X 2 (t) = G. This is so because ( F G) Λ = ( ) A BR 1 B T Q A }{{ T } Hamiltonian Matrix - H ( F G). (42) so that P = G dx 1 1 dt + dx 2 F dt F 1 G = GF 1 FΛF 1 + GΛF 1 = 0. M.E. University of Minnesota 249
39 This proposition shows that there are Cn 2n (i.e. 2n!/n!) solutions of P, depending on which n of the 2n eigenvectors of H are picked to define F and G. PropositionSuppose that (A, B) is stabilizable and (A, C) is detectable. Then, the eigenvalues of H are symmetrically located across the imaginary and real axes with no eigenvalues on the imaginary axis. Proof: Consider a invertible coordinate transformation T = ( ) I 0, T 1 = P I n ( ) I 0. P I n Hence, T 1 HT = A BR 1 B T }{{ P } BR 1 B T A c 0 (A BR 1 B T P ) }{{ T } A T c Since T 1 HT and H share the same eigenvalues, this shows that H contains the eigenvalues of A c as well M.E. University of Minnesota 250
40 as of A T c. Hence, n eigenvalues of H must lie on the closed RHP, and n eigenvalues lie on the closed LHP. In other words, the eigenvalues of H are symmetrically located about both the real and imaginary axes. Further, A c (and hence H) cannot have any eigenvalues on the imaginary axis. For, otherwise, the optimal cost will be infinite. Hence, H must have n eigenvalues on the open LHP, and n on the open RHP. Since we know that the closed loop system matrix: A c = A BR 1 B T P = A BR 1 B T GF 1. must be stable if (A, B) is stabilizable and (A, C) is detectable, we have the following result. Proposition Suppose that (A, B) is stabilizable and (A, C) is detectable. The steady state ( solution of the F CTRDE is the P = GF 1 where are chosen G) to consist of the n eigenvectors that correspond to the stable eigenvalues of H. M.E. University of Minnesota 251
41 Proof: Since A c = A BR 1 B T P = A BR 1 B T GF 1 ( A c F = [A, BR 1 B T F ] = FΛ G) where the last equality is obtained from Eq.(42). Hence, diag(λ) consists of the eigenvalues of A c, and columns of F are the eigenvectors. Since (A, B) and (A, C) are stabilizable and detectable, A c is stable. Thus, Λ must have negative real parts. Remark Integrating the Hamiltonian matrix is not a good idea, either in forward time or in reverse time, since either way will be unstable. Integrating the Riccati backwards in time is more reliable. The Hamiltonian matrix is useful for solving for the solution to the ARE though, via its eigenvalues and eigenvectors. M.E. University of Minnesota 252
42 Selection of Q and R The quality of the control design depends on the choice of Q and R (and for finite time S also). How should one choose these? Some suggestions here, most taken from (Anderson and Moore, 1990). Generally an iterative design/simulation process is needed; If there is a specific output z = Cx that need to be kept small, choose Q = C T C. Use physically meaningful state and control variables and use physical insights to select Q and R. Choose Q and R to be diagonal in the absence of information about coupling. Obtain acceptable excursions: x i (t) x i,max, u i (t) u i,max, x i (t f ) x i,f max Then choose Q, R and S to be inversely proportional to x 2 i,max, u2 i,max and x2 i,f max respectively. M.E. University of Minnesota 253
43 Off diagonal terms in Q reflect coupling. e.g. to coordinate( x 1 = kx ) 2, one can choose C = [1 k] so 1 k that Q = k k 2. One can add other objectives to Q. For finite time regulator problem with time interval T. The ratio of the running cost objective and the terminal cost objective should be scaled by 1/T and the dimension of x R n and u R m : tf =t 0 +T t 0 1 nt xt Qx+ 1 mt ut Ru dt+x T (t f )Sx(t f ). where Q, R, S are selected based on separate x(t), u(t) and x(t f ) criteria. Additional relative scalings should be iteratively determined. If R = diag[r 1, r 2, r 3 ] and after simulation, u 2 is too large, increase r 2 ; If after simulation, state x 3 is too large, modify Q such that x T Qx x T Qx + γx 2 3 etc. If performance is related to frequency, use frequency weighting (see below). M.E. University of Minnesota 254
44 Summary from Infinite Horizon LQ Performance criteria ẋ = Ax + Bu; x(t 0 ) = x 0. J(u,x 0, t 0, t f ) = 1 2 Solution: tf t 0 [ x T (t)qx(t) + u T (t)ru(t) ] dt P(t) = A T P(t) + P(t)A P(t)BR 1 B T (t)p + Q(t); P(t f ) = 0. If (A, B) is controllable (or stabilizable), then P = P(t ) (the same as P(t 0 ) as t f ) exists. The optimal control is: u o (t) = R 1 B T }{{ P } x(t). K and the cost performance is: J o (x 0 ) = 1 2 xt 0 P x 0. M.E. University of Minnesota 255
45 Furthermore, let Q = C T C. If (A, C) is observable (or detectable), then: is stable. A BK Thus, if (A, B) controllable or at least stabilizable and with suitable choice of R - must be positive definite Q = C T C - (A, C) should be observable or at least detectable then LQ control methodology automatically generates a feedback gain K such that A BK is stable. Solution of ARE can be generated by solving the eigenvalues / eigenvectors of the 2n 2n Hamiltonian matrix in Eq.(34). The eigenvalues are symmetrically located w.r.t. both the real and imaginary axis. The true solution (i.e. P = P(t f )) is the one associated with the stable eigenvalues of the Hamiltonian matrix. M.E. University of Minnesota 256
46 LQ Regulator for Discrete Time Systems Consider the discrete time system: x(k + 1) = A(k)x(k) + B(k)u(k); x(0) = x 0. with the performance criteria given by: J(u( ), x 0 ) = 1 2 xt (k f )Sx(k f ) k f 1 [ x T (k)qx(k) + u T (k)ru(k) ]. k=k 0 The optimal control is given by: u o (k) = K(k) x(k) K(k) = [R(k) + B T (k)p(k + 1)B(k)] 1 B T (k)p(k + 1)A(k) where P(k) is the solution to the discrete time Riccati difference equation: P(k) =Q(k) + A T (k)p(k + 1)A(k) A T P(k + 1)B(k) [ R + B T (k)p(k + 1)B(k) ] 1 B T (k)p(k + 1)A(k); P(k f ) = S. M.E. University of Minnesota 257
47 The optimal cost-to-go at time k is: J o (x(k), k) = 1 2 xt (k)p(k)x(k). Notice that positive definiteness condition on R(k) implies that: R(k) + B T (k)p(k + 1)B(k) is invertible. Exercise: Derive the discrete time LQ result using dynamic programming. M.E. University of Minnesota 258
48 Discrete time LTI LQR For A, B, Q and R being constants, we have: P(k ) P satisfies the Algebraic Discrete Time Riccatti Equation: A T P A A T P B[R+B T P B] 1 B T PA+Q = P If (A, B) is controllable and (A, C) (where Q = C T C) is observable, then P is the unique positive definite solution. If (A, C) is only detectable, then P is the positive semi-definite solution. The feedback gain is then: K = [R + B T P B] 1 B T P A The closed loop system is stable, meaning that all eigenvalues of A BK have magnitudes less than 1 (lie in the unit disk centered at the origin). M.E. University of Minnesota 259
49 Other LQ topics - see notes for details Infinite LQ design methodology ensures a set of gains such that the closed loop poles are on the open LHP for continuous time system, and inside a unit disk for a discrete time system. The above properties can be exploited to ensure that closed loop poles lie in a certain region (left of α or within a disk). Frequency weighting can be used to penalized control or performance. Approach is to design weighting filters, and then convert the Frequency Shaped LQ into a standard LQ problem. The LQ gains satisfy a so-called Return Difference Equality, from which robustness properties can be derived. Asymptotic closed loop pole locations as r (expensive control) or r 0 (cheap control) can be derived using the Symmetric Root Locus (a consequence of the return difference equality). The optimal cost-to-go is: J o (x,k) = x T P(k)x. M.E. University of Minnesota 260
50 Eigenvalue placements LQR can be thought of as a way of generating stabilizing feedback gains. However, exactly where the closed loop poles are in the LHP is not clear. We now propose a couple of ways in which we can exert some control over them. The idea is to transform the problem. In this section, we assume that (A, B) is controllable, and (A, C) is observable where Q = C T C. M.E. University of Minnesota 261
51 Guaranteed convergence rate To move the poles so that they are at least to the left of α (i.e. if the eigenvalues of A BK are λ i, we want Re(λ i ) < α, hence more stable), we solve an alternate problem. Since ẋ = A x + Bu Re(eig(A BK)) < 0 Thus, setting A = A + αi, we solve the LQ problem for the plant: ẋ = (A + αi)x + Bu. This ensures that the eigenvalues of Re((A + αi) BK) < 0. Notice that (A + αi) BK and A BK have the same eigenvectors. Thus, the eigenvalues of A BK, say λ i and those of A + αi BK, σ i, are related by λ i = σ i α. Since Re(σ i ) < 0, Re(λ i ) < α. M.E. University of Minnesota 262
52 Eigenvalues to lie in a disk A more interesting case is to ensure that the eigenvalues of the closed loop system lie in a disk centered at ( α,0) and with radius ρ < α. This, in addition to specifying the convergence rate to be faster than α ρ, it also specifies limits for the damping ratio, so that the system will not be too oscillatory. The idea is to use the discrete time LQ solution, which ensures that the eigenvalues of A BK lie in a unit disk centered at the origin. We need to scale the disk and to translate it. Let the continuous time plant be: ẋ = Ax + Bu If we solve the discrete time LQ problem for the plant, x(k + 1) = 1 ρ A x(k) + 1 ρ Bu(k) then, the eigenvalues of 1 ρ (A BK) would lie in the unit disk and the eigenvalues of (A BK) would lie in the disk with radius ρ, both centered at the origin. M.E. University of Minnesota 263
53 Using the same trick as before, we now translate the eigenvalues by α by setting A = A + αi. In summary, if we use the discrete time LQ control design method for the plant x(k + 1) = 1 ρ (A + αi)x(k) + 1 ρ Bu(k) then, the eigenvalues of 1 ρ ((A + αi) BK) would lie within the unit disk centered at the origin. This implies that the eigenvalues of ((A + αi) BK) lie in a disk of radius ρ centered at the origin. Finally, this implies that the eigenvalues of A BK lie in a disk or radius ρ centered at ( α,0). M.E. University of Minnesota 264
54 Frequency Shaping Original LQ problem is specified in the time domain. The cost function is the L 2 norms of the control, and of z = Q 1 2x. Frequency domain sometimes more useful. example For In dual stage actuator, one actuator prefers large amplitude low frequency, the other prefers high frequency small amplitude disturbances lie within a narrow bandwidth Robustness are easier to specify in the frequency domain (e.g. in loop shaping concepts) Parseval theorem For a squared integrable function h(t) R p with ht (t)h(t)dt <, h T (t)h(t)dt = 1 2π H (jw)h(jw)dw (43) where H(jw) is the fourier transform or as H(s = jw), i.e. the Laplace transform of h(t) evaluated at s = jw. H (jw) denotes the conjugate transpose M.E. University of Minnesota 265
55 of H(jw). Hence, for H(s) with real coefficient, H (jw) = H( jw) T. Parseval theorem states that the energy (L 2 norm) in the signal can be evaluated either in the frequency or in the time domain. So, suppose that we want to optimize the criteria in the frequency domain as: J(u) = 1 2π X (jw)q 1(jw)Q 1 (jw)x(jw) + U (jw)r 1(jw)R 1 (jw)u(jw) dw (44) This says that the state and control weightings are given by Q(w 2 ) = Q 1(jw)Q 1 (jw); R(w 2 ) = R 1(jw)R 1 (jw). If we define X 1 (jw) = Q 1 (jw)x(jw), U 1 (jw) = R 1 (jw)u(jw), then J(u) = 1 2π X 1(jw)X 1 (jw) + U 1(jw)U 1 (jw) dw M.E. University of Minnesota 266
56 Now, apply Parseval Theorem in reverse, J(u) = x T 1 (t)x 1 (t) + u T 1 (t)u 1 (t) dt. (45) If we know the dynamics of x 1 and u 1 is the control input, then we can solve using the standard LQ technique. We express the filters Q 1 (s) and R 1 (s) as filters (e.g. low pass and high pass) with the actual state and input of the system x(t) and u(t) as inputs, and frequency weighted state x 1 (t) and u 1 (t) as outputs: Q 1 (s) = C Q (si A Q ) 1 B Q + D Q (46) R 1 (s) = C R (si A R ) 1 B R + D R (47) which says that in the time domain: and similarly, ż 1 = A Q z 1 + B Q x (48) x 1 = C Q z 1 + D Q x (49) ż 2 = A R z 2 + B R u (50) u 1 = C R z 2 + D R u. (51) M.E. University of Minnesota 267
57 Hence we can define an augmented plant: d x z 1 = A 0 0 B Q A Q 0 x z 1 + B 0 dt 0 0 A R z 2 z 2 B R u(t) or with x = [x;z 1 ; z 2 ], etc. x = Ā x + Bu. Since u 1 = ( 0 0 C R ) x + DR u x 1 = ( D Q C Q 0 ) x the cost function Eq.(45) becomes: ) J(u) = ( x T u T)( Q e N )( x T dt (52) N R e u where Q e = DQ TD Q DQ TC Q 0 C Q TD Q CQ TC Q CR TC R M.E. University of Minnesota 268
58 N = 0 0 ; R e = DRD T R. CR TD R Eq.(52) is still not in standard form yet because of the off diagonal block N. We can convert Eq.(52) into the standard form if we consider: u(t) = R 1 e N x + v (53) The integrand in Eq.(52) becomes: )( )( ) T ( x v T)( I N T Re 1 Qe N T I 0 0 I N R e Re 1 N I)( x v = ( x ) T v T)( Q e N T Re 1 N 0 0 R e)( x v Then, define Q = Q e N T R T e N, R = Re (54) and new state dynamics: x = (Ā BR 1 e N) x + Bv (55) M.E. University of Minnesota 269
59 and cost function, J(v) = x T Q x + v T Rv dt. (56) Eqs.(55)-(56) are then in the standard LQ format. The stabilizability and detectability conditions are now needed for the the augmented system (what are they?). M.E. University of Minnesota 270
Extensions and applications of LQ
Extensions and applications of LQ 1 Discrete time systems 2 Assigning closed loop pole location 3 Frequency shaping LQ Regulator for Discrete Time Systems Consider the discrete time system: x(k + 1) =
More informationLinear Quadratic Optimal Control
156 c Perry Y.Li Chapter 6 Linear Quadratic Optimal Control 6.1 Introduction In previous lectures, we discussed the design of state feedback controllers using using eigenvalue (pole) placement algorithms.
More informationOPTIMAL CONTROL. Sadegh Bolouki. Lecture slides for ECE 515. University of Illinois, Urbana-Champaign. Fall S. Bolouki (UIUC) 1 / 28
OPTIMAL CONTROL Sadegh Bolouki Lecture slides for ECE 515 University of Illinois, Urbana-Champaign Fall 2016 S. Bolouki (UIUC) 1 / 28 (Example from Optimal Control Theory, Kirk) Objective: To get from
More informationAdvanced Mechatronics Engineering
Advanced Mechatronics Engineering German University in Cairo 21 December, 2013 Outline Necessary conditions for optimal input Example Linear regulator problem Example Necessary conditions for optimal input
More informationOptimal Control. Quadratic Functions. Single variable quadratic function: Multi-variable quadratic function:
Optimal Control Control design based on pole-placement has non unique solutions Best locations for eigenvalues are sometimes difficult to determine Linear Quadratic LQ) Optimal control minimizes a quadratic
More informationSteady State Kalman Filter
Steady State Kalman Filter Infinite Horizon LQ Control: ẋ = Ax + Bu R positive definite, Q = Q T 2Q 1 2. (A, B) stabilizable, (A, Q 1 2) detectable. Solve for the positive (semi-) definite P in the ARE:
More informationQuadratic Stability of Dynamical Systems. Raktim Bhattacharya Aerospace Engineering, Texas A&M University
.. Quadratic Stability of Dynamical Systems Raktim Bhattacharya Aerospace Engineering, Texas A&M University Quadratic Lyapunov Functions Quadratic Stability Dynamical system is quadratically stable if
More informationLecture 4 Continuous time linear quadratic regulator
EE363 Winter 2008-09 Lecture 4 Continuous time linear quadratic regulator continuous-time LQR problem dynamic programming solution Hamiltonian system and two point boundary value problem infinite horizon
More informationEN Applied Optimal Control Lecture 8: Dynamic Programming October 10, 2018
EN530.603 Applied Optimal Control Lecture 8: Dynamic Programming October 0, 08 Lecturer: Marin Kobilarov Dynamic Programming (DP) is conerned with the computation of an optimal policy, i.e. an optimal
More informationTopic # Feedback Control Systems
Topic #17 16.31 Feedback Control Systems Deterministic LQR Optimal control and the Riccati equation Weight Selection Fall 2007 16.31 17 1 Linear Quadratic Regulator (LQR) Have seen the solutions to the
More informationRobotics. Control Theory. Marc Toussaint U Stuttgart
Robotics Control Theory Topics in control theory, optimal control, HJB equation, infinite horizon case, Linear-Quadratic optimal control, Riccati equations (differential, algebraic, discrete-time), controllability,
More information1. Find the solution of the following uncontrolled linear system. 2 α 1 1
Appendix B Revision Problems 1. Find the solution of the following uncontrolled linear system 0 1 1 ẋ = x, x(0) =. 2 3 1 Class test, August 1998 2. Given the linear system described by 2 α 1 1 ẋ = x +
More informationState Regulator. Advanced Control. design of controllers using pole placement and LQ design rules
Advanced Control State Regulator Scope design of controllers using pole placement and LQ design rules Keywords pole placement, optimal control, LQ regulator, weighting matrixes Prerequisites Contact state
More informationLinear-Quadratic Optimal Control: Full-State Feedback
Chapter 4 Linear-Quadratic Optimal Control: Full-State Feedback 1 Linear quadratic optimization is a basic method for designing controllers for linear (and often nonlinear) dynamical systems and is actually
More information6. Linear Quadratic Regulator Control
EE635 - Control System Theory 6. Linear Quadratic Regulator Control Jitkomut Songsiri algebraic Riccati Equation (ARE) infinite-time LQR (continuous) Hamiltonian matrix gain margin of LQR 6-1 Algebraic
More informationEE363 homework 2 solutions
EE363 Prof. S. Boyd EE363 homework 2 solutions. Derivative of matrix inverse. Suppose that X : R R n n, and that X(t is invertible. Show that ( d d dt X(t = X(t dt X(t X(t. Hint: differentiate X(tX(t =
More informationMODERN CONTROL DESIGN
CHAPTER 8 MODERN CONTROL DESIGN The classical design techniques of Chapters 6 and 7 are based on the root-locus and frequency response that utilize only the plant output for feedback with a dynamic controller
More informationLecture 9: Discrete-Time Linear Quadratic Regulator Finite-Horizon Case
Lecture 9: Discrete-Time Linear Quadratic Regulator Finite-Horizon Case Dr. Burak Demirel Faculty of Electrical Engineering and Information Technology, University of Paderborn December 15, 2015 2 Previous
More informationMATH4406 (Control Theory) Unit 6: The Linear Quadratic Regulator (LQR) and Model Predictive Control (MPC) Prepared by Yoni Nazarathy, Artem
MATH4406 (Control Theory) Unit 6: The Linear Quadratic Regulator (LQR) and Model Predictive Control (MPC) Prepared by Yoni Nazarathy, Artem Pulemotov, September 12, 2012 Unit Outline Goal 1: Outline linear
More informationStochastic and Adaptive Optimal Control
Stochastic and Adaptive Optimal Control Robert Stengel Optimal Control and Estimation, MAE 546 Princeton University, 2018! Nonlinear systems with random inputs and perfect measurements! Stochastic neighboring-optimal
More information9 Controller Discretization
9 Controller Discretization In most applications, a control system is implemented in a digital fashion on a computer. This implies that the measurements that are supplied to the control system must be
More information6.241 Dynamic Systems and Control
6.241 Dynamic Systems and Control Lecture 24: H2 Synthesis Emilio Frazzoli Aeronautics and Astronautics Massachusetts Institute of Technology May 4, 2011 E. Frazzoli (MIT) Lecture 24: H 2 Synthesis May
More informationLQR, Kalman Filter, and LQG. Postgraduate Course, M.Sc. Electrical Engineering Department College of Engineering University of Salahaddin
LQR, Kalman Filter, and LQG Postgraduate Course, M.Sc. Electrical Engineering Department College of Engineering University of Salahaddin May 2015 Linear Quadratic Regulator (LQR) Consider a linear system
More informationSuppose that we have a specific single stage dynamic system governed by the following equation:
Dynamic Optimisation Discrete Dynamic Systems A single stage example Suppose that we have a specific single stage dynamic system governed by the following equation: x 1 = ax 0 + bu 0, x 0 = x i (1) where
More informationMath Ordinary Differential Equations
Math 411 - Ordinary Differential Equations Review Notes - 1 1 - Basic Theory A first order ordinary differential equation has the form x = f(t, x) (11) Here x = dx/dt Given an initial data x(t 0 ) = x
More informationHere represents the impulse (or delta) function. is an diagonal matrix of intensities, and is an diagonal matrix of intensities.
19 KALMAN FILTER 19.1 Introduction In the previous section, we derived the linear quadratic regulator as an optimal solution for the fullstate feedback control problem. The inherent assumption was that
More informationFormula Sheet for Optimal Control
Formula Sheet for Optimal Control Division of Optimization and Systems Theory Royal Institute of Technology 144 Stockholm, Sweden 23 December 1, 29 1 Dynamic Programming 11 Discrete Dynamic Programming
More informationProblem 1 Cost of an Infinite Horizon LQR
THE UNIVERSITY OF TEXAS AT SAN ANTONIO EE 5243 INTRODUCTION TO CYBER-PHYSICAL SYSTEMS H O M E W O R K # 5 Ahmad F. Taha October 12, 215 Homework Instructions: 1. Type your solutions in the LATEX homework
More informationModule 05 Introduction to Optimal Control
Module 05 Introduction to Optimal Control Ahmad F. Taha EE 5243: Introduction to Cyber-Physical Systems Email: ahmad.taha@utsa.edu Webpage: http://engineering.utsa.edu/ taha/index.html October 8, 2015
More informationTheorem 1. ẋ = Ax is globally exponentially stable (GES) iff A is Hurwitz (i.e., max(re(σ(a))) < 0).
Linear Systems Notes Lecture Proposition. A M n (R) is positive definite iff all nested minors are greater than or equal to zero. n Proof. ( ): Positive definite iff λ i >. Let det(a) = λj and H = {x D
More informationTime-Invariant Linear Quadratic Regulators!
Time-Invariant Linear Quadratic Regulators Robert Stengel Optimal Control and Estimation MAE 546 Princeton University, 17 Asymptotic approach from time-varying to constant gains Elimination of cross weighting
More informationOptimal control and estimation
Automatic Control 2 Optimal control and estimation Prof. Alberto Bemporad University of Trento Academic year 2010-2011 Prof. Alberto Bemporad (University of Trento) Automatic Control 2 Academic year 2010-2011
More information5. Observer-based Controller Design
EE635 - Control System Theory 5. Observer-based Controller Design Jitkomut Songsiri state feedback pole-placement design regulation and tracking state observer feedback observer design LQR and LQG 5-1
More informationLinear System Theory
Linear System Theory Wonhee Kim Chapter 6: Controllability & Observability Chapter 7: Minimal Realizations May 2, 217 1 / 31 Recap State space equation Linear Algebra Solutions of LTI and LTV system Stability
More informationModeling and Analysis of Dynamic Systems
Modeling and Analysis of Dynamic Systems Dr. Guillaume Ducard Fall 2017 Institute for Dynamic Systems and Control ETH Zurich, Switzerland G. Ducard c 1 / 57 Outline 1 Lecture 13: Linear System - Stability
More informationCONTROL DESIGN FOR SET POINT TRACKING
Chapter 5 CONTROL DESIGN FOR SET POINT TRACKING In this chapter, we extend the pole placement, observer-based output feedback design to solve tracking problems. By tracking we mean that the output is commanded
More informationESC794: Special Topics: Model Predictive Control
ESC794: Special Topics: Model Predictive Control Discrete-Time Systems Hanz Richter, Professor Mechanical Engineering Department Cleveland State University Discrete-Time vs. Sampled-Data Systems A continuous-time
More informationCourse Outline. Higher Order Poles: Example. Higher Order Poles. Amme 3500 : System Dynamics & Control. State Space Design. 1 G(s) = s(s + 2)(s +10)
Amme 35 : System Dynamics Control State Space Design Course Outline Week Date Content Assignment Notes 1 1 Mar Introduction 2 8 Mar Frequency Domain Modelling 3 15 Mar Transient Performance and the s-plane
More informationModule 07 Controllability and Controller Design of Dynamical LTI Systems
Module 07 Controllability and Controller Design of Dynamical LTI Systems Ahmad F. Taha EE 5143: Linear Systems and Control Email: ahmad.taha@utsa.edu Webpage: http://engineering.utsa.edu/ataha October
More informationTheory in Model Predictive Control :" Constraint Satisfaction and Stability!
Theory in Model Predictive Control :" Constraint Satisfaction and Stability Colin Jones, Melanie Zeilinger Automatic Control Laboratory, EPFL Example: Cessna Citation Aircraft Linearized continuous-time
More informationNumerical Optimal Control Overview. Moritz Diehl
Numerical Optimal Control Overview Moritz Diehl Simplified Optimal Control Problem in ODE path constraints h(x, u) 0 initial value x0 states x(t) terminal constraint r(x(t )) 0 controls u(t) 0 t T minimize
More informationLECTURE NOTES IN CALCULUS OF VARIATIONS AND OPTIMAL CONTROL MSc in Systems and Control. Dr George Halikias
Ver.1.2 LECTURE NOTES IN CALCULUS OF VARIATIONS AND OPTIMAL CONTROL MSc in Systems and Control Dr George Halikias EEIE, School of Engineering and Mathematical Sciences, City University 4 March 27 1. Calculus
More information2 The Linear Quadratic Regulator (LQR)
2 The Linear Quadratic Regulator (LQR) Problem: Compute a state feedback controller u(t) = Kx(t) that stabilizes the closed loop system and minimizes J := 0 x(t) T Qx(t)+u(t) T Ru(t)dt where x and u are
More informationECSE.6440 MIDTERM EXAM Solution Optimal Control. Assigned: February 26, 2004 Due: 12:00 pm, March 4, 2004
ECSE.6440 MIDTERM EXAM Solution Optimal Control Assigned: February 26, 2004 Due: 12:00 pm, March 4, 2004 This is a take home exam. It is essential to SHOW ALL STEPS IN YOUR WORK. In NO circumstance is
More informationTopic # Feedback Control
Topic #5 6.3 Feedback Control State-Space Systems Full-state Feedback Control How do we change the poles of the state-space system? Or,evenifwecanchangethepolelocations. Where do we put the poles? Linear
More informationME Fall 2001, Fall 2002, Spring I/O Stability. Preliminaries: Vector and function norms
I/O Stability Preliminaries: Vector and function norms 1. Sup norms are used for vectors for simplicity: x = max i x i. Other norms are also okay 2. Induced matrix norms: let A R n n, (i stands for induced)
More informationControl Systems I. Lecture 4: Diagonalization, Modal Analysis, Intro to Feedback. Readings: Emilio Frazzoli
Control Systems I Lecture 4: Diagonalization, Modal Analysis, Intro to Feedback Readings: Emilio Frazzoli Institute for Dynamic Systems and Control D-MAVT ETH Zürich October 13, 2017 E. Frazzoli (ETH)
More informationModule 02 CPS Background: Linear Systems Preliminaries
Module 02 CPS Background: Linear Systems Preliminaries Ahmad F. Taha EE 5243: Introduction to Cyber-Physical Systems Email: ahmad.taha@utsa.edu Webpage: http://engineering.utsa.edu/ taha/index.html August
More informationPrinciples of Optimal Control Spring 2008
MIT OpenCourseWare http://ocw.mit.edu 16.323 Principles of Optimal Control Spring 2008 For information about citing these materials or our Terms of Use, visit: http://ocw.mit.edu/terms. 16.323 Lecture
More informationExam. 135 minutes, 15 minutes reading time
Exam August 6, 208 Control Systems II (5-0590-00) Dr. Jacopo Tani Exam Exam Duration: 35 minutes, 5 minutes reading time Number of Problems: 35 Number of Points: 47 Permitted aids: 0 pages (5 sheets) A4.
More informationEE221A Linear System Theory Final Exam
EE221A Linear System Theory Final Exam Professor C. Tomlin Department of Electrical Engineering and Computer Sciences, UC Berkeley Fall 2016 12/16/16, 8-11am Your answers must be supported by analysis,
More informationLinear Quadratic Regulator (LQR) Design I
Lecture 7 Linear Quadratic Regulator LQR) Design I Dr. Radhakant Padhi Asst. Proessor Dept. o Aerospace Engineering Indian Institute o Science - Bangalore LQR Design: Problem Objective o drive the state
More informationRobust Control 5 Nominal Controller Design Continued
Robust Control 5 Nominal Controller Design Continued Harry G. Kwatny Department of Mechanical Engineering & Mechanics Drexel University 4/14/2003 Outline he LQR Problem A Generalization to LQR Min-Max
More informationChapter III. Stability of Linear Systems
1 Chapter III Stability of Linear Systems 1. Stability and state transition matrix 2. Time-varying (non-autonomous) systems 3. Time-invariant systems 1 STABILITY AND STATE TRANSITION MATRIX 2 In this chapter,
More information4F3 - Predictive Control
4F3 Predictive Control - Discrete-time systems p. 1/30 4F3 - Predictive Control Discrete-time State Space Control Theory For reference only Jan Maciejowski jmm@eng.cam.ac.uk 4F3 Predictive Control - Discrete-time
More informationEE C128 / ME C134 Feedback Control Systems
EE C128 / ME C134 Feedback Control Systems Lecture Additional Material Introduction to Model Predictive Control Maximilian Balandat Department of Electrical Engineering & Computer Science University of
More informationOutline. 1 Linear Quadratic Problem. 2 Constraints. 3 Dynamic Programming Solution. 4 The Infinite Horizon LQ Problem.
Model Predictive Control Short Course Regulation James B. Rawlings Michael J. Risbeck Nishith R. Patel Department of Chemical and Biological Engineering Copyright c 217 by James B. Rawlings Outline 1 Linear
More informationNonlinear Observers. Jaime A. Moreno. Eléctrica y Computación Instituto de Ingeniería Universidad Nacional Autónoma de México
Nonlinear Observers Jaime A. Moreno JMorenoP@ii.unam.mx Eléctrica y Computación Instituto de Ingeniería Universidad Nacional Autónoma de México XVI Congreso Latinoamericano de Control Automático October
More informationTime-Invariant Linear Quadratic Regulators Robert Stengel Optimal Control and Estimation MAE 546 Princeton University, 2015
Time-Invariant Linear Quadratic Regulators Robert Stengel Optimal Control and Estimation MAE 546 Princeton University, 15 Asymptotic approach from time-varying to constant gains Elimination of cross weighting
More information16.31 Fall 2005 Lecture Presentation Mon 31-Oct-05 ver 1.1
16.31 Fall 2005 Lecture Presentation Mon 31-Oct-05 ver 1.1 Charles P. Coleman October 31, 2005 1 / 40 : Controllability Tests Observability Tests LEARNING OUTCOMES: Perform controllability tests Perform
More informationHomework Solution # 3
ECSE 644 Optimal Control Feb, 4 Due: Feb 17, 4 (Tuesday) Homework Solution # 3 1 (5%) Consider the discrete nonlinear control system in Homework # For the optimal control and trajectory that you have found
More informationLecture 2: Discrete-time Linear Quadratic Optimal Control
ME 33, U Berkeley, Spring 04 Xu hen Lecture : Discrete-time Linear Quadratic Optimal ontrol Big picture Example onvergence of finite-time LQ solutions Big picture previously: dynamic programming and finite-horizon
More informationChap. 3. Controlled Systems, Controllability
Chap. 3. Controlled Systems, Controllability 1. Controllability of Linear Systems 1.1. Kalman s Criterion Consider the linear system ẋ = Ax + Bu where x R n : state vector and u R m : input vector. A :
More informationLecture 2 and 3: Controllability of DT-LTI systems
1 Lecture 2 and 3: Controllability of DT-LTI systems Spring 2013 - EE 194, Advanced Control (Prof Khan) January 23 (Wed) and 28 (Mon), 2013 I LTI SYSTEMS Recall that continuous-time LTI systems can be
More informationThe goal of this chapter is to study linear systems of ordinary differential equations: dt,..., dx ) T
1 1 Linear Systems The goal of this chapter is to study linear systems of ordinary differential equations: ẋ = Ax, x(0) = x 0, (1) where x R n, A is an n n matrix and ẋ = dx ( dt = dx1 dt,..., dx ) T n.
More informationEL2520 Control Theory and Practice
EL2520 Control Theory and Practice Lecture 8: Linear quadratic control Mikael Johansson School of Electrical Engineering KTH, Stockholm, Sweden Linear quadratic control Allows to compute the controller
More informationME 234, Lyapunov and Riccati Problems. 1. This problem is to recall some facts and formulae you already know. e Aτ BB e A τ dτ
ME 234, Lyapunov and Riccati Problems. This problem is to recall some facts and formulae you already know. (a) Let A and B be matrices of appropriate dimension. Show that (A, B) is controllable if and
More informationNonlinear Control Lecture 5: Stability Analysis II
Nonlinear Control Lecture 5: Stability Analysis II Farzaneh Abdollahi Department of Electrical Engineering Amirkabir University of Technology Fall 2010 Farzaneh Abdollahi Nonlinear Control Lecture 5 1/41
More informationLINEAR-CONVEX CONTROL AND DUALITY
1 LINEAR-CONVEX CONTROL AND DUALITY R.T. Rockafellar Department of Mathematics, University of Washington Seattle, WA 98195-4350, USA Email: rtr@math.washington.edu R. Goebel 3518 NE 42 St., Seattle, WA
More information1. The Transition Matrix (Hint: Recall that the solution to the linear equation ẋ = Ax + Bu is
ECE 55, Fall 2007 Problem Set #4 Solution The Transition Matrix (Hint: Recall that the solution to the linear equation ẋ Ax + Bu is x(t) e A(t ) x( ) + e A(t τ) Bu(τ)dτ () This formula is extremely important
More informationDynamic Optimal Control!
Dynamic Optimal Control! Robert Stengel! Robotics and Intelligent Systems MAE 345, Princeton University, 2017 Learning Objectives Examples of cost functions Necessary conditions for optimality Calculation
More informationOn Irregular Linear Quadratic Control: Deterministic Case
1 On Irregular Linear Quadratic Control: Deterministic Case Huanshui Zhang and Juanjuan Xu arxiv:1711.9213v2 math.oc 1 Mar 218 Abstract This paper is concerned with fundamental optimal linear quadratic
More informationECE7850 Lecture 7. Discrete Time Optimal Control and Dynamic Programming
ECE7850 Lecture 7 Discrete Time Optimal Control and Dynamic Programming Discrete Time Optimal control Problems Short Introduction to Dynamic Programming Connection to Stabilization Problems 1 DT nonlinear
More informationOPTIMAL CONTROL SYSTEMS
SYSTEMS MIN-MAX Harry G. Kwatny Department of Mechanical Engineering & Mechanics Drexel University OUTLINE MIN-MAX CONTROL Problem Definition HJB Equation Example GAME THEORY Differential Games Isaacs
More informationME8281-Advanced Control Systems Design
ME8281 - Advanced Control Systems Design Spring 2016 Perry Y. Li Department of Mechanical Engineering University of Minnesota Spring 2016 Lecture 4 - Outline 1 Homework 1 to be posted by tonight 2 Transition
More information4F3 - Predictive Control
4F3 Predictive Control - Lecture 2 p 1/23 4F3 - Predictive Control Lecture 2 - Unconstrained Predictive Control Jan Maciejowski jmm@engcamacuk 4F3 Predictive Control - Lecture 2 p 2/23 References Predictive
More informationA Globally Stabilizing Receding Horizon Controller for Neutrally Stable Linear Systems with Input Constraints 1
A Globally Stabilizing Receding Horizon Controller for Neutrally Stable Linear Systems with Input Constraints 1 Ali Jadbabaie, Claudio De Persis, and Tae-Woong Yoon 2 Department of Electrical Engineering
More informationAustralian Journal of Basic and Applied Sciences, 3(4): , 2009 ISSN Modern Control Design of Power System
Australian Journal of Basic and Applied Sciences, 3(4): 4267-4273, 29 ISSN 99-878 Modern Control Design of Power System Atef Saleh Othman Al-Mashakbeh Tafila Technical University, Electrical Engineering
More information5 Handling Constraints
5 Handling Constraints Engineering design optimization problems are very rarely unconstrained. Moreover, the constraints that appear in these problems are typically nonlinear. This motivates our interest
More informationSUCCESSIVE POLE SHIFTING USING SAMPLED-DATA LQ REGULATORS. Sigeru Omatu
SUCCESSIVE POLE SHIFING USING SAMPLED-DAA LQ REGULAORS oru Fujinaka Sigeru Omatu Graduate School of Engineering, Osaka Prefecture University, 1-1 Gakuen-cho, Sakai, 599-8531 Japan Abstract: Design of sampled-data
More informationDeterministic Dynamic Programming
Deterministic Dynamic Programming 1 Value Function Consider the following optimal control problem in Mayer s form: V (t 0, x 0 ) = inf u U J(t 1, x(t 1 )) (1) subject to ẋ(t) = f(t, x(t), u(t)), x(t 0
More informationControl Systems. Design of State Feedback Control.
Control Systems Design of State Feedback Control chibum@seoultech.ac.kr Outline Design of State feedback control Dominant pole design Symmetric root locus (linear quadratic regulation) 2 Selection of closed-loop
More informationEE C128 / ME C134 Final Exam Fall 2014
EE C128 / ME C134 Final Exam Fall 2014 December 19, 2014 Your PRINTED FULL NAME Your STUDENT ID NUMBER Number of additional sheets 1. No computers, no tablets, no connected device (phone etc.) 2. Pocket
More informationDissipativity. Outline. Motivation. Dissipative Systems. M. Sami Fadali EBME Dept., UNR
Dissipativity M. Sami Fadali EBME Dept., UNR 1 Outline Differential storage functions. QSR Dissipativity. Algebraic conditions for dissipativity. Stability of dissipative systems. Feedback Interconnections
More informationMathematical Economics. Lecture Notes (in extracts)
Prof. Dr. Frank Werner Faculty of Mathematics Institute of Mathematical Optimization (IMO) http://math.uni-magdeburg.de/ werner/math-ec-new.html Mathematical Economics Lecture Notes (in extracts) Winter
More informationLecture 5 Linear Quadratic Stochastic Control
EE363 Winter 2008-09 Lecture 5 Linear Quadratic Stochastic Control linear-quadratic stochastic control problem solution via dynamic programming 5 1 Linear stochastic system linear dynamical system, over
More informationPrinciples of Optimal Control Spring 2008
MIT OpenCourseWare http://ocw.mit.edu 16.323 Principles of Optimal Control Spring 2008 For information about citing these materials or our Terms of Use, visit: http://ocw.mit.edu/terms. 16.323 Lecture
More informationChapter 3. LQ, LQG and Control System Design. Dutch Institute of Systems and Control
Chapter 3 LQ, LQG and Control System H 2 Design Overview LQ optimization state feedback LQG optimization output feedback H 2 optimization non-stochastic version of LQG Application to feedback system design
More informationLyapunov Stability Theory
Lyapunov Stability Theory Peter Al Hokayem and Eduardo Gallestey March 16, 2015 1 Introduction In this lecture we consider the stability of equilibrium points of autonomous nonlinear systems, both in continuous
More informationZeros and zero dynamics
CHAPTER 4 Zeros and zero dynamics 41 Zero dynamics for SISO systems Consider a linear system defined by a strictly proper scalar transfer function that does not have any common zero and pole: g(s) =α p(s)
More informationEML5311 Lyapunov Stability & Robust Control Design
EML5311 Lyapunov Stability & Robust Control Design 1 Lyapunov Stability criterion In Robust control design of nonlinear uncertain systems, stability theory plays an important role in engineering systems.
More informationMASSACHUSETTS INSTITUTE OF TECHNOLOGY Department of Electrical Engineering and Computer Science : Dynamic Systems Spring 2011
MASSACHUSETTS INSTITUTE OF TECHNOLOGY Department of Electrical Engineering and Computer Science 6.4: Dynamic Systems Spring Homework Solutions Exercise 3. a) We are given the single input LTI system: [
More informationFirst-Order Low-Pass Filter!
Filters, Cost Functions, and Controller Structures! Robert Stengel! Optimal Control and Estimation MAE 546! Princeton University, 217!! Dynamic systems as low-pass filters!! Frequency response of dynamic
More informationIntroduction to Modern Control MT 2016
CDT Autonomous and Intelligent Machines & Systems Introduction to Modern Control MT 2016 Alessandro Abate Lecture 2 First-order ordinary differential equations (ODE) Solution of a linear ODE Hints to nonlinear
More informationOptimization-Based Control. Richard M. Murray Control and Dynamical Systems California Institute of Technology
Optimization-Based Control Richard M. Murray Control and Dynamical Systems California Institute of Technology Version v2.1b (2 Oct 218) c California Institute of Technology All rights reserved. This manuscript
More informationDirect Methods. Moritz Diehl. Optimization in Engineering Center (OPTEC) and Electrical Engineering Department (ESAT) K.U.
Direct Methods Moritz Diehl Optimization in Engineering Center (OPTEC) and Electrical Engineering Department (ESAT) K.U. Leuven Belgium Overview Direct Single Shooting Direct Collocation Direct Multiple
More informationLinear System Theory. Wonhee Kim Lecture 1. March 7, 2018
Linear System Theory Wonhee Kim Lecture 1 March 7, 2018 1 / 22 Overview Course Information Prerequisites Course Outline What is Control Engineering? Examples of Control Systems Structure of Control Systems
More informationSubject: Optimal Control Assignment-1 (Related to Lecture notes 1-10)
Subject: Optimal Control Assignment- (Related to Lecture notes -). Design a oil mug, shown in fig., to hold as much oil possible. The height and radius of the mug should not be more than 6cm. The mug must
More informationECEEN 5448 Fall 2011 Homework #5 Solutions
ECEEN 5448 Fall 211 Homework #5 Solutions Professor David G. Meyer December 8, 211 1. Consider the 1-dimensional time-varying linear system ẋ t (u x) (a) Find the state-transition matrix, Φ(t, τ). Here
More informationStochastic optimal control theory
Stochastic optimal control theory Bert Kappen SNN Radboud University Nijmegen the Netherlands July 5, 2008 Bert Kappen Introduction Optimal control theory: Optimize sum of a path cost and end cost. Result
More information