MATRIX Motor with Individual Control Capability for Iron Loss Suppression under Flux Weakening Control

Transcription

1 IEEJ Journal of Industry Applications Vol.5 No. pp DOI: /ieejjia.5.17 Paper MATRIX Motor with Individual Control Capability for Iron Loss Suppression under Flux Weakening Control Hiroki Hijikata Student Member, Kan Akatsu Member Yoshihiro Miyama Member, Hideaki Arita Member Akihiro Daikoku Member Manuscript received Aug. 8, 014, revised Nov. 19, 015) Interior type permanent magnet synchronous motors IPMSMs) have been widely used for various applications because of their high efficiency and high power density. It is well known that the flux weakening control of an IPMSM can extend the operating range by reducing the back-emf and producing reluctance torque. An IPMSM, however, experiences a large harmonic iron loss in the stator teeth caused by the distorted flux density. In particular, the harmonic eddy current loss significantly increases in the high speed region under flux weakening control. This paper describes a novel motor-drive system that includes an individual multi-phase drive in order to control the current and voltage in each armature winding. The proposed system can be expected to achievehigh efficiency driving, a wide operating range, fail-safety and so on. Furthermore, the proposed method gives the distorted flux density distribution a sinusoidal waveform, because the individual control can observe and regulate the instantaneous flux linkage in each stator tooth, although the motor consists of a full-pitch distributed winding. Therefore, the proposed drive can suppress the harmonic iron loss under the flux weakening operation. Simulation and experimental results show that the proposed method can achieve an extensive reduction in the total iron loss in particular in the high speed region. Keywords: harmonic iron loss, individual control, interior permanent magnet synchronous motor, MATRIX motor, multi-phase drive 1. Introduction Recently, the power semiconductor switching devices for the motor-drive application have been greatly developed as the low on-state resistance, low switching losses, high power density, and high frequency operation. In particular, it is said that some switching devices, such as SJ-MOSFET, already exceeded the theoretical limitation 1). Some literatures have investigated the motor-drive system with multiple power semiconductor switching devices ) 5). Although these researches require more number of power electronics devices, the motors realize the additional advantages. For example, the expansion method of the driving range is proposed by using the open winding connected with the dual inverters configuration ). The electronic winding change method is also researched in order to extend the operating range ). The pole change technique with multi-phase inverter is studied in a case of induction motor for variable speed operation 4) 5). The authors have previously proposed the MATRIX motor which is the concept of a new integrated motor drive system ). As the additional advantage, it was shown that the motor can vary the operating range and can achieve high Shibaura Institute of Technology -7-5, Toyosu, Koto-ku, Tokyo , Japan Mitsubishi Electric Corporation, Advanced Technology R&D Center 8-1-1, Tsukaguchi-Honmachi, Amagasaki, Hyogo 1-81, Japan efficiency driving under various conditions as requested by the arbitrary winding connection. The motor configuration, however, uses the ineffective armature winding for producing the output torque even though the main advantage of the integrated system is high power density. The ineffective winding decreases the utilization ratio of the armature winding which leads low winding factor. In addition, the previous motor has low degree of freedom due to the three phase inverter and conventional wye-connection. To overcome these problems, this paper proposes a novel motor-drive system which includes the multi-phase open-end windings in order to individually control the current and voltage in each armature winding. The armature windings are connected with the insulated H-bridge inverters for the arbitrary exciting waveform. The configuration can obtain variable magnet flux linkage and variable inductance. Therefore, the proposed scheme can improve the utilization ratio of the armature winding keeping with the achievement of the variable parameters. An interior type permanent magnet synchronous motor IPMSM) can extend the operating range facilitated by the flux weakening control 7). It is popularly known that the control method can reduce the back-emf and can generate the reluctance torque. On the other hand, the IPMSM experiences a large harmonic iron loss at high rotational speed under the flux weakening operation 8) 9). In particular, the harmonic eddy current loss is significantly increased in the stator teeth of the distributed wound IPMSM 10) 11). Since the flux weakening control can only cancel the fundamental c 01 The Institute of Electrical Engineers of Japan. 17

2 component of the magnet flux linkage, the cause of increasing iron loss is the harmonic magnetomotive forces by the permanent magnet and armature reaction. Some literatures investigate the optimized design methods of the rotor or stator core to reduce the harmonic component 1) 15). These methods, however, also reduce the output torque due to decreasing magnet flux linkage and q-axis inductance. As a result, these structural approaches take difficult to achieve high efficiency in especially low speed region due to the degraded output torque. This paper presents a novel suppression control method for the harmonic iron loss by individual winding current control. The proposed method makes the distorted flux density distribution to sinusoidal waveform in each stator tooth even though the motor consists of a full-pitch distributed winding. Since the proposed method has the multiple open-end winding connected with the individual H-bridge inverters, the multi-phase drive which obtains the additional degree of freedom is possible. It is known that the fundamental component and harmonic component, such as rd and 5 th order, can be individually controlled by the remaining degree of freedom in the multi-phase motor 1) 17). In the proposed drive system, it is possible to inject the harmonic component to compensate the harmonic iron loss by decoupling current control in the fundamental component, rd order subspace, and 5 th order subspace. In this paper, the proposed motor drive system is firstly described. Section introduces the prototype model which is the IPMSM with 8-pole and 48-slot. The section also shows the decoupling transformation matrix of the prototype model. Section reveals that the proposed control method can particularly reduce harmonic iron loss in FEA. Finally, Sect. 4 presents the experimental results of the proposed control method.. Proposed Motor Drive Description.1 Motor and Inverter Configurations Figure 1a) shows the proposed principle motor which is down sized to the lab. scale. The model which is the IPMSM with the fullpitch distributed windings includes the open-winding either wye- or delta-connection ) 18). Although the open-winding motor requires more number of power electronics devices than the number of the conventional wye-connection uses, it is known that the configuration is expected to offer additional advantages, for example, utilizing the zero-sequence, the high utilization ratio of DC bus voltage, and its fault tolerance capability with an individual drive unit. The armature windings connected with H-bridge inverters are shown in Fig. 1b). The motor require -phase armature windings due to the flux control in each tooth. The model specifications are summarized in Table 1.. Definition of Frame Frames which are used in this paper are shown in Fig.. The d 1 -q 1 frame is rotational orthogonal frame which is synchronized with the rotor angle. Where, i dq1, θ and β are the current vector of the fundamental component, electric angle, and current phase, respectively.. Decoupling Transformation Matrix Since the output torque is stemmed from vector control produced by the interaction of the fundamental field component and fundamental stator current component, the conventional -phase motor with Y connection requires only two stator current frames d-q current component). On the other hand, the multi-phase motor can obtain additional degree of freedom. For example, it is previously investigated that the current and voltage vector of the -phase motor with open-end winding can be evolved by subspace of the original six-dimensional space 1) 17). Of course two components of the six must be the fundamental component which is electromechanical energy conversion function of the electric motor. The rest of the four components are harmonic component, such as rd and 5 th order subspace. In the -phase drive, the following equation is used to transform -phase stationary coordinate system into six rotational reference frames. xdq = T x phase 1) x phase = T 1 x dq ) Table 1. Specification of principle motor a) Motor structure b) Inverters and armature windings Fig. 1. Proposed motor drive system. The motor has the 8-pole 48-slot configuration of the IPMSM with the single-layer full-pitch distributed-winding configuration Fig.. Winding arrangement of the proposed motor and definition of frame. Two sets of three phase windings spatially by 0 electric degrees 177 IEEJ Journal IA, Vol.5, No., 01

3 xdq = xd1 x q1 x d x q x d5 x q5 T x phase = xa x B x C x D x E x F T where x is the sampled data which can be either current or voltage. x dq andx phase are the sampled vector in the rotational frame and stationary frame, respectively. T is defined as the decoupling transformation matrix to control the harmonics and given by ) at the bottom of this page 1) 17). Each row in ) is defined as following. First and second rows are the fundamental component of the machine variables and the n 1 th order harmonics with n 1 = 1m 1 ± 1 where m 1 is 1,,..., such as 11 th,1 th, rd,5 th... harmonics, which are transformed into the d 1 -q 1 subspace or d 1 - q 1 plane. Third and fourth rows in ) show n th order harmonics with n = m where m = 1,, 5..., such as rd,9 th,15 th,1 st...harmonics, which mapped into the d -q subspace or d -q plane. Last two rows in ) indicate n 5 th order harmonics with n 5 = m 5 ± 1 where m 5 = 1,, 5..., such as 5 th,7 th,17 th,19 th...harmonics, which mapped into the d 5 -q 5 subspace or d 5 -q 5 plane. The fundamental component, d -q subspace, and d 5 -q 5 subspace are individually controlled since these subspaces are orthogonal each other. Since there are not the mutual between fundamental component, d -q subspace and d 5 -q 5 subspace, each component is extracted by followings in order to control the individual components xdqn = Tn x phase 4) xdqn = xdn x qn T where x dqn isn th sampled vector in the rotational frame. n is time harmonic order n = 1,, or 5). T n is the transformation matrix shown in 5) at the bottom of this page..4 Teeth Flux Control to Suppress Iron Loss The flux linkage is estimated by following equation 19) λ = v Ri) dt+λ 0 ) where, λ, v, R, i, andλ 0 denote the total flux linkage, armature voltage, armature winding resistance, armature current, and initial value of the total flux linkage, respectively. Although the system requires voltage sensors and current sensors, it can obtain the flux linkage in each tooth by λ Teeth = C λ phase 7) λ Teeth = T λ AB λ BC λ CD λ DE λ EF λ FA C = 8) T λ phase = λa λ B λ C λ D λ E λ F where, λ Teeth andλ phase indicate the total flux linkage in the teeth and in the armature winding, respectively. C is the conversion matrix. Using ) and 8), the flux linkage expresses in the rotational orthogonal frame. The difference between the reference of the flux linkage and distorted flux linkage produce the fluctuated voltage reference in order to cancel the distorted flux linkage. Figure shows the block diagram to suppress the iron loss. The proposed method consists of two parts which are current control and teeth flux control. The part of current control regulates the fundamental component to produce the output torque. The part of teeth flux control uses the harmonic components which are remaining degree of freedom in the T = cos θ sin θ cos θ π ) sin θ π ) cos θ cos θ π ) sin θ sin θ π ) cos 5θ cos 5 θ π ) sin 5θ sin 5 θ π ) cos θ π ) sin θ π ) cos θ π ) sin θ π ) cos 5 θ π ) sin 5 θ π ) cos θ π ) sin θ π ) cos θ π ) sin θ π ) cos 5 θ π ) sin 5 θ π ) cos θ π ) sin θ π ) cos θ π ) sin θ π ) cos 5 θ π ) sin 5 θ π ) cos θ 5π ) sin θ 5π ) cos θ 5π ) sin θ 5π ) cos 5 θ 5π ) sin 5 θ 5π ) ) T n = cos nθ sin nθ cos n θ π ) sin n θ π ) cos n θ π ) sin n θ π ) cos n θ π ) sin n θ π ) cos n θ π ) sin n θ π ) cos n θ 5π ) sin n θ 5π ) 5) 178 IEEJ Journal IA, Vol.5, No., 01

4 Fig.. Block diagram of proposed control method -phase. The fundamental component is extracted by 4) and 5) when 1 is substituted into n. The current controller is PI control. In case of the teeth flux control, the subspace of d -q and d 5 -q 5 components are extracted by 4) and 5) when or 5 are substituted into n. The flux estimator ) and 7) calculates the flux linkage in each tooth. The input voltage reference is modified by the flux controller which establishes the alternative voltage reference. The flux controller is P control. As mentioned above, there are not the mutual between fundamental component, d -q subspace and d 5 -q 5 subspace. The previous study also discussed about eliminating the cross coupling effects in dual -phase motor drive 17). The decoupling control can be done by same method of the conventional -phase motor drive. The above method makes the harmonic component of flux density in each tooth reduced by additional harmonic current even though the fundamental component does not change. Thus, the harmonic iron losses caused by distorted flux density can be suppressed.. Simulation Results and Discussion This section confirms the effectiveness of the suppression method for the iron loss by the FEA. In this verification, a coupled analysis using the FEA JMAG from JSOL Corporation) and circuit simulator MATLAB Simulink from Math- Works Corporation) is carried out. The applying current is used by the ideal voltage source in order to compare the iron loss without the iron loss caused by PWM switching..1 Iron Loss Calculation In this paper, the FEA calculates the total iron loss by following expressions with the Fourier transformation of the flux density 9) W i = K e D s nf) {B r,n + B θ,n }dv iron + iron n n K h D s nf) {B r,n + B θ,n }dv 9) where, W i, K e, K h, D s, f,andv denote the total iron loss, coefficient of the eddy current loss, coefficient of the hysteresis loss, density of the electrical steel plate, frequency, and volume of the motor core, respectively. The value of K e and K h are and 11., respectively. B r,n and B θ,n indicate the n th harmonics of the radial and tangential components of the flux density. The total iron loss depends on the amplitude of flux density and frequency. Therefore, it is said that the iron loss rapidly increases with the increasing harmonic a) Flux density distribution b) FFT results Fig. 4. Radial component of flux density distribution and FFT results at point I in Fig. 1a). The applying phase current is 7.5 A rms and current phase β shifts from 0 to 80 deg. component of the flux density.. Distorted Flux Density under Flux Weakening Control It is generally known that the flux weakening control distorts the flux density waveform in stator teeth 1). This part shows the FEA results of the flux density distribution under the conventional flux weakening control. Figure 4 shows the radial component of the flux density distribution waveforms at point I in Fig. 1a) and their FFT results. The applying phase current is 7.5 A rms and current phase shifts from 0 to 80 deg. As shown in these graphs, it is found that the rd,5 th,7 th,and9 th harmonics component of the flux density significantly increase with increasing current phase even though the fundamental component decreases. Figure 5 shows the FEA results of the classified iron loss in case of driving at the low and high speed region. The conditions of the phase current and rotating speed are 7.5 A rms and 100 or 000 rpm, respectively. These values in Fig. 5 are normalized by each total iron loss at 0 deg. As shown in this graph, the total iron losses decrease with increasing current angle in each rotating speed. In case of the low speed conditions, it seems that the total iron loss is occupied by the hysteresis loss in the stator core due to low frequency in 9). On the other hand, it can be found that the eddy current losses in the high speed conditions, especially in stator core, are much larger than those in the low speed conditions. In particular, these losses cannot be decreased with increasing current angle because the harmonic eddy current loss caused by the distorted flux density in the stator teeth remarkably increases.. Comparison of Conventional Flux Weakening Control and Proposed Control This section describes a comparison between the conventional flux weakening control and proposed iron loss suppression control. The simulation conditions are summarized in Table. Figure shows the waveforms with or without compensation and Fig. 7 shows their FFT results. As shown in Figs. a) and 7a), the flux density at point I is much distorted without compensation which is the conventional flux weakening control, especially rd,5 th,7 th,9 th harmonics. In contrast, the flux density with compensation which is the proposed method has few harmonic components. The proposed control method can achieve the flux density deformed to substantially sinusoidal waveform. The voltage waveform also include few harmonic component with compensation compared with the 179 IEEJ Journal IA, Vol.5, No., 01

5 Fig. 5. Classified iron loss in case of the low and high speed operation. The applying phase current is 7.5 A rms and current phase shifts from 0 to 80 deg. These values are normalized by each total iron loss at 0 deg Table. Simulation conditions a) Flux density distribution at point I b) Phase current of A winding a) Flux density distribution at point I b) Phase current of A winding c) Phase voltage of A winding d) Output torque Fig. 7. FFT results of each waveform c) Phase voltage of A winding d) Output torque Fig.. FEA results of each waveform with or without compensation conventional method in Figs. c) and 7c). It is found that the peak voltage with compensation can be reduced. Thus, the proposed method has capability for expanding operating range. On the other hand, the conventional method applies only the fundamental current, whereas the proposed method makes the harmonic current to flow in Figs. b) and 7b). The harmonic current is provided to cancel the rd,5 th,7 th, and 9 th harmonics of the flux density. As shown in Figs. d) and 7d), the output torque of the proposed control method is pulsated because the applying harmonic current causes the large torque ripple. However, it can be said that the torque ripples less effect in the high-speed region when the flux weakening control is performed..4 Iron Loss Comparison This part compares the iron loss with or without compensation. Figure 8 shows the iron loss density distribution with or without compensation. As shown in this figure, the iron loss in the stator teeth is clearly reduced with compensation. It can be said that the proposed method can particularly suppress the iron loss a) Without compensation b) With compensation Fig. 8. Iron loss density distribution in the rotor and stator core because the flux density waveform in each tooth becomes sinusoidal waveform as shown in Figs. a) and 7a). Thus, the proposed control method can largely inhibit the expression of the harmonic iron loss. Figures 9 and 10 show the results of the classified iron losses due to their origins and the ratio of the output power and loss analysis, respectively. It can be seen that the stator core eddy current loss with compensation is remarkably decreased in Fig. 9. Furthermore, other losses are also decreased such as the stator core hysteresis loss, rotor core hysteresis loss, and rotor core eddy current loss. The total iron loss decreases approximately 70% whereas the copper loss 180 IEEJ Journal IA, Vol.5, No., 01

6 Fig. 11. Picture of experimental equipment Fig. 9. Classified iron losses with or without compensation Table. Experimental conditions Fig. 10. Output power and loss abalysis. Each value in the bar is expressed in watt W) increases about 5% compared with the conventional flux weakening control due to applying additional harmonic current in Fig. 10. The efficiency is improved approximately 1% even though the copper loss is largely dependent on the total loss. Therefore, it is confirmed that the proposed method can be effect especially for a high proportion machine of the iron loss in the high speed region. 4. Experimental Results and Discussion 4.1 Experimental Conditions This part experimentally confirms the effectiveness of the proposed control method compared with conventional method. The manufactured motor structure is processed strictly according to FEA model in Fig. 1a). The illustration of the experimental description is shown in Fig. 11. The experimental conditions are shown in Table. In this part, the rotational speed is sifted from 1000 rpm to 4000 rpm due to the experimental limitation. 4. Experimental Verification This part shows the experimental results of the proposed control method compared with conventional method. The waveforms of the measured current and observed flux linkage and their FFT results are shown in Figs. 1 and 1, respectively when the motor rotates in 4000 rpm. As shown in Figs. 1a) and 1a), it is found that the observed flux linkage waveform in stator teeth without compensation which is the conventional method involves much harmonic component as with the flux density waveform in FEA results. It is assumed that the radial component of the flux density also gets distorted. On the other hand, the harmonic component with compensation which is the proposed method became very small. The waveform of the flux linkage became nearly sinusoidal. In particular, the proposed method causes a significant decrease in the rd,5 th, a) Flux linkage in stator teeth between A and B winding b) Phase current of A winding Fig. 1. Experimental results of each waveform with or without compensation a) Flux linkage in stator teeth between A and B winding b) Phase current of A winding Fig. 1. FFT results of each experimental waveform with or without compensation 7 th and 9 th harmonic components. It is said that the proposed control method can achieve the flux linkage deformed to substantially sinusoidal waveform. In case of applying current in Figs. 1b) and 1b), the conventional method makes the current to approximately sinusoidal waveform by current control. In contrast, the proposed method allows the current to distorted waveform as with the FEA results because the distorted reference is used for flux control to suppress the harmonic components. It is found that the same components as the flux linkage, such as rd,5 th,7 th,and9 th harmonics, is applied in order to deform the flux linkage. Figure 14 shows the relationship between rotating speed and iron loss in both FEA and experiment with and without compensation. In each rotating speed, the output torque is constant. The experimental results measured by the power analyzer WT 1800 Yokogawa electric corporation). The iron loss is obtained by the subtraction of the output power and copper loss which is calculated by RMS value of the applying 181 IEEJ Journal IA, Vol.5, No., 01

7 Fig. 14. Comparison of the iron loss with and without compensation in each rotational speed current and winding resistance from the input power. In experimental results, the different of the iron loss in two cases is much larger than FEA results because the measured iron losses in experiment include the mechanical loss. On the other hand, it is found that the measured iron loss can be surely reduced when the proposed compensation method is applied. In the particularly high speed region, the measured iron loss is significantly reduced as is the case in FEA results. Therefore, it is experimentally verified that the proposed control method can suppress the iron loss. 5. Conclusion This research proposed the novel motor-drive system which includes the individual multi-phase drive in order to control the current and voltage in each armature winding. This paper described the suppression control method for the harmonic iron loss of the IPMSM under the flux weakening control. The results of this study reached the following conclusions. The proposed method can regulate each tooth flux utilizing the individual control. As a result, the harmonic iron losses are reduced at the high rotational speed by the harmonic current injection. Some simulation results reveal that the proposed control method obtains the effectiveness in the high speed region. In particular, the iron loss can be reduced up to about 70% even though the copper loss increases about 5% in 000 rpm. The experimental results confirmed that the proposed method can reduce the harmonic flux linkage in the stator teeth. Furthermore, it is found that the iron loss is surely reduced when the proposed compensation method is applied. References 1 ) A. Nakagawa: Theoretical Investigation of Silicon Limit Characteristics of IGBT, in Proc. of IEEE 18th International Symposium on Power Semiconductor Devices & IC s, pp ) ) D. Pan, F. Liang, Y. Wang, and T.A. 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Appl., Vol., No.5, pp ) Appendix Symbols A, B, C, D, E, F Phase name of -phase β Current phase B r,n, B θ,n n th harmonics of radial and tangential components of flux density C Conversion matrix D s Density of electrical steel plate f frequency i Armature current K e, K h Coefficient of eddy current loss and hysteresis loss λ Total flux linkage λ θ Initial value of total flux linkage λ Teeth Total flux linkage in stator teeth n, n 1, n, n 5 Harmonic order θ Electric angle 18 IEEJ Journal IA, Vol.5, No., 01

8 R Resistance of armature winding T Transformation matrix T n n th order transformation matrix v Volume of motor core x Sampled data that is either current or flux x phase Sampled vector in stationary frame x dq Sampled vector in rotational frame x dqn n th order sampled vector in rotational frame Total iron loss W i Hiroki Hijikata Student Member) was born in Saitama, Japan in He received B.E. and M.E. degree in electrical engineering from Shibaura Institute of Technology, Tokyo, Japan in 011 and 01, respectively. He is currently working toward the Ph.D. degree. His research interests are electric motor drive especially for multi-phase machine. Mr. Hijikata is a student member of IEEE. Yoshihiro Miyama Member) was born in Hyogo, Japan in 198. He received the B.E. and M.E. degrees in electrical and electronic engineering from Ritsumeikan University, Shiga, Japan in 007 and 009, respectively. He joined Advanced Technology R&D Center of Mitsubishi Electric Corporation. Mr. Miyama is a member of the IEEE IAS and IEE of Japan. Hideaki Arita Member) was born in Oita, Japan in He received the B.E. and M.E. degrees in electrical and electronic engineering from Oita University, Oita, Japan in 000 and 00, respectively. He joined Advanced Technology R&D Center of Mitsubishi Electric Corporation. Kan Akatsu Member) received B.S., M.S., and Ph.D. degrees in electrical engineering from Yokohama National University, Yokohama, Japan, in 1995, 1997, 000 respectively. He joined Nissan Research Center, Yokosuka, Japan, in 000, he contributed to the design and analysis of the new concept permanent magnet machines. In 00, he joined the department of Electrical and Electric Engineering at Tokyo University of Agriculture and Technology, Tokyo, Japan, as an assistant professor. From 005 to 007, he is a JSPS Postdoctoral Fellowship for Research Abroad, visiting professor in WEM- PEC Wisconsin Electric Machines and Power Electronics Consortium), University of Wisconsin-Madison. In 009, he is an Associate Professor with the department of Electrical Engineering at Shibaura Institute of Technology, Tokyo, Japan, where he is currently a Full Professor. His research interests are motor control, motor design and inverter control. Dr. Akatsu is a member of the IEEE PELS, IAS, IE and IEE of Japan. Akihiro Daikoku Member) was born in Osaka, Japan in 195. He received the B.E., M.E., and Ph.D. degrees in electrical engineering from Kyoto University, Kyoto, Japan in 1989, 1991 and 007, respectively. He joined Advanced Technology R&D Center of Mitsubishi Electric Corporation in 1991, and since 015 he is a senior manager of the Electromechanical Technology Department in the corporation. Dr. Daikoku is a member of the IEEE and IEE of Japan. 18 IEEJ Journal IA, Vol.5, No., 01