Achievable Rates and Optimal Resource Allocation for Imperfectly-Known Fading Relay Channels

Transcription

1 Achievable Rates and Optimal Resource Allocation for Imperfectly-Known Fading Relay Channels Junwei Zhang, Mustafa Cenk Gursoy Department of Electrical Engineering arxiv:83.992v [cs.it] 3 Mar 28 University of Nebraska-Lincoln, Lincoln, NE jzhang3@bigred.unl.edu, gursoy@engr.unl.edu Abstract In this paper, achievable rates and optimal resource allocation strategies for imperfectly-known fading relay channels are studied. It is assumed that communication starts with the network training phase in which the receivers estimate the fading coefficients of their respective channels. In the data transmission phase, amplify-and-forward and decode-and-forward relaying schemes with different degrees of cooperation are considered, and the corresponding achievable rate expressions are obtained. Three resource allocation problems are addressed: ) power allocation between data and training symbols; 2) time/bandwidth allocation to the relay; 3) power allocation between the source and relay in the presence of total power constraints. The achievable rate expressions are employed to identify the optimal resource allocation strategies. Finally, energy efficiency is investigated by studying the bit energy requirements in the low-snr regime. Index Terms: Relay channel, cooperative transmission, channel estimation, imperfectly-known fading channels, achievable rates, optimal resource allocation, energy efficiency in the low-power regime. I. INTRODUCTION In wireless communications, deterioration in performance is experienced due to various impediments such as interference, fluctuations in power due to reflections and attenuation, and randomly-varying channel conditions caused by mobility and changing environment. Recently, cooperative wireless communications has attracted much interest as a technique that can mitigate these degradations and provide higher rates or improve the reliability through diversity gains. The relay channel was first introduced by van der Meulen in [], and initial research was primarily conducted to understand the rates achieved in relay channels [2], This work was supported in part by the NSF CAREER Grant CCF The material in this paper was presented in part at the 45th Annual Allerton Conference on Communication, Control and Computing in Sept. 27. March 3, 28 DRAFT

2 [3]. More recently, diversity gains of cooperative transmission techniques have been studied in [4] [7]. In [6], several cooperative protocols have been proposed, with amplify-and-forward (AF) and decode-andforward (DF) being the two basic relaying schemes. The performance of these protocols are characterized in terms of outage events and outage probabilities. In [8], three different time-division AF and DF cooperative protocols with different the degrees of broadcasting and receive collision are studied. In general, the area has seen an explosive growth in the number of studies (see e.g., [9], [], [], [2], [3], and references therein). An excellent review of cooperative strategies from both rate and diversity improvement perspectives is provided in [4] in which the impacts of cooperative schemes on device architecture and higher-layer wireless networking protocols are also addressed. Recently, a special issue has been dedicated to models, theory, and codes for relaying and cooperation in communication networks in [5]. As noted above, studies on relaying and cooperation are numerous. However, most work has assumed that the channel conditions are perfectly known at the receiver and/or transmitter sides. Especially in mobile applications, this assumption is unwarranted as randomly-varying channel conditions can be learned by the receivers only imperfectly. Moreover, the performance analysis of cooperative schemes in such scenarios is especially interesting and called for because relaying introduces additional channels and hence increases uncertainty in the model if the channels are known only imperfectly. Recently, Wang et al. in [6] considered pilot-assisted transmission over wireless sensory relay networks, and analyzed scaling laws achieved by the amplify-and-forward scheme in the asymptotic regimes of large nodes, large block length, and small SNR values. In this study, the channel conditions are being learned only by the relay nodes. In [7], Avestimehr and Tse studied the outage capacity of slow fading relay channels. They showed that Bursty Amplify-Forward strategy achieves the outage capacity in the low SNR and low outage probability regime. Interestingly, they further proved that the optimality of Bursty AF is preserved even if the receivers do not have prior knowledge of the channels. In this paper, we study the achievable rates of imperfectly-known fading relay channels. We assume that transmission takes place in two phases: network training phase and data transmission phase. In the network training phase, a priori unknown fading coefficients are estimated at the receivers with the assistance of pilot symbols. Following the training phase, AF and DF relaying techniques with different degrees of cooperation are employed in the data transmission. We first obtain achievable rate expressions for different relaying protocols and subsequently identify optimal resource allocation strategies that maximize the rates. We consider three types of resource allocation problems: ) power allocation between data and training 2

3 symbols; 2) time/bandwidth allocation to the relay; 3) power allocation between the source and relay if there is a total power constraint in the system. Finally, we investigate the energy efficiency by finding the bit energy requirements in the low-snr regime. The organization of the rest of the paper is as follows. In Section II, we describe the channel model. Network training and data transmission phases are explained in Section III. We obtain the achievable rate expressions in Section IV and study the optimal resource allocation strategies in Section V. We discuss the energy efficiency in the low-snr regime in Section VI. Finally, we provide conclusions in Section VII. II. CHANNEL MODEL We consider the three-node relay network which consists of a source, destination, and a relay node. x s Source h sr Relay y r x r h rd h sd y d r y d Destination Source-destination, source-relay, and relay-destination channels are modeled as Rayleigh block-fading channels with fading coefficients denoted by h sr, h sd, and h rd, respectively for each channel. Due to the blockfading assumption, the fading coefficients h sr CN(, σ 2 sr ), h sd CN(, σ 2 sd ), and h rd CN(, σ 2 rd ) 2 stay constant for a block of m symbols before they assume independent realizations for the following block. In this system, the source node tries to send information to the destination node with the help of the intermediate relay node. It is assumed that the source, relay, and destination nodes do not have prior knowledge of the realizations of the fading coefficients. The transmission is conducted in two phases: network training phase in which the fading coefficients are estimated at the receivers, and data transmission phase. Overall, the source and relay are subject to the following power constraints in one block: x s,t 2 + E{ x s 2 } mp s, () x r,t 2 + E{ x r 2 } mp r. (2) where x s,t and x r,t are the source and relay training signal vectors, respectively, and x s and x r are the corresponding source and relay data vectors. 2 x CN(d,σ 2 ) is used to denote a proper complex Gaussian random variable with mean d and variance σ 2. 3

4 III. NETWORK TRAINING AND DATA TRANSMISSION A. Network Training Phase Each block transmission starts with the training phase. In the first symbol period, source transmits a pilot symbol to enable the relay and destination to estimate the channel coefficients h sr and h sd, respectively. In the average power limited case, sending a single pilot is optimal because instead of increasing the number of pilot symbols, a single pilot with higher power can be used. The signals received by the relay and destination are y r,t = h sr x s,t + n r, and y d,t = h sd x s,t + n d, (3) respectively. Similarly, in the second symbol period, relay transmits a pilot symbol to enable the destination to estimate the channel coefficient h rd. The signal received by the destination is y r d,t = h rdx r,t + n r d. (4) In the above formulations, n r CN(, N ), n d CN(, N ), and n r d CN(, N ) represent independent Gaussian noise samples at the relay and the destination nodes. In the training process, it is assumed that the receivers employ minimum mean-square-error (MMSE) estimation. We assume that the source allocates δ s of its total power for training while the relay allocates δ r of its total power for training. As described in [25], the MMSE estimate of h sr is given by ĥ sr = σ2 sr δs mp s y σsrδ 2 r,t, (5) s mp s + N where y r,t CN(, σ 2 sr δ smp s + N ). We denote by h sr the estimate error which is a zero-mean complex Gaussian random variable with variance var( h sr ) = and h rd, we have σ 2 sr N σ 2 sr δsmps+n. Similarly, for the fading coefficients h sd ĥ sd = ĥ rd = σ2 sd δs mp s σsd 2 δ y d,t, y d,t CN(, σsd 2 smp s + N δ smp s + N ), var( h sd ) = σ2 rd δr mp r σrd 2 δ yd,t r rmp r + N, yr d,t CN(, σ2 rd δ rmp r + N ), var( h rd ) = σ 2 sd N σ 2 sd δ smp s + N, (6) σ 2 rd N σ 2 rd δ rmp r + N. (7) With these estimates, the fading coefficients can now be expressed as h sr = ĥsr + h sr, h sd = ĥsd + h sd, h rd = ĥrd + h rd. (8) 4

5 B. Data Transmission Phase The practical relay node usually cannot transmit and receive data simultaneously. Thus, we assume that the relay works under half-duplex constraint. Hence, the relay first listens and then transmits. As discussed in the previous section, within a block of m symbols, the first two symbols are allocated to network training. In the remaining duration of m 2 symbols, data transmission takes place. We introduce the relay transmission parameter α and assume that α(m 2) symbols are allocated for relay transmission. Hence, α can be seen as the fraction of total time or bandwidth allocated to the relay. Note that the parameter α enables us to control the degree of cooperation. We consider several transmission protocols which can be classified into two categories by whether or not the source and relay simultaneously transmits information: non-overlapped and overlapped transmission. Note that in both cases, the relay transmits over a duration of α(m 2) symbols. In non-overlapped transmission, source transmits over a duration of ( α)(m 2) symbols and becomes silent as the relay transmits. On the other hand, in overlapped transmission, source transmits all the time and sends m 2 symbols in each block. We assume that the data vectors x s and x r are composed of independent random variables with equal energy. Hence, the covariance matrices of x s are given by E{x s x s } = P s I = ( δ s)mp s (m 2)( α) I, and E{x sx s } = P s2 I = ( δ s)mp s (m 2) in non-overlapped and overlapped transmissions, respectively. The covariance matrix for x r is I, (9) E{x r x r } = P r I = ( δ r)mp r (m 2)α I. () ) Non-overlapped transmission: We first consider the two simplest cooperative protocols: non-overlapped AF, and non-overlapped DF with repetition coding where the relay decodes the message and re-encodes it using the same codebook as the source. In these protocols, since the relay either amplifies the received signal, or decodes it but uses the same codebook as the source when forwarding, source and relay should be allocated equal time slots in the cooperation phase. Therefore, we initially have direct transmission from the source to the destination without any aid from the relay over a duration of ( 2α)(m 2) symbols. In this phase, source sends x s and the received signal at the destination is given by y d = h sd x s + n d. () 5

6 Subsequently, cooperative transmission starts. At first, the source transmits an α(m 2)-dimensional symbol vector x s2 which is received at the the relay and the destination, respectively, as y r = h sr x s2 + n r, and y d2 = h sd x s2 + n d2. (2) For compact representation, we denote the signal received at the destination directly from the source by y d = [yd T yt d2 ]T where T denotes the transpose operation. Next, the source becomes silent, and the relay transmits an α(m 2)-dimensional symbol vector x r which is generated from the previously received y r [6] [7]. This approach corresponds to protocol 2 in [8], which realizes the maximum degrees of broadcasting and exhibits no receive collision. The destination receives y r d = h rdx r + n r d. (3) After substituting the expressions in (8) into () (3), we have y d = ĥsdx s + h sd x s + n d, y r = ĥsrx s2 + h sr x s2 + n r, y d2 = ĥsdx s2 + h sd x s2 + n d2, (4) y r d = ĥrdx r + h rd x r + n r d. (5) We define the source data vector as x s = [x T s x T s2] T. Note that we have < α /2 for AF and repetition coding DF. Therefore, α = /2 models full cooperation while we have noncooperative communications as α. It should also be noted that α should in general be chosen such that α(m 2) is an integer. For non-overlapped transmission, we also consider DF with parallel channel coding, in which the relay uses a different codebook to encode the message. In this case, the source and relay do not have to be allocated the same duration in the cooperation phase. Therefore, source transmits over a duration of ( α)(m 2) symbols while the relay transmits in the remaining duration of α(m 2) symbols. Clearly, the range of α is now < α <. In this case, the input-output relations are given by (2) and (3). Since there is no separate direct transmission, x s2 = x s and y d2 = y d in (2). Moreover, the dimensions of the vectors x s,y d,y r are now ( α)(m 2), while x r and yd r are vectors of dimension α(m 2). 2) Overlapped transmission: In this category, we consider a more general and complicated scenario in which the source transmits all the time. In AF and repetition DF, similarly as in the non-overlapped model, cooperative transmission takes place in the duration of 2α(m 2) symbols. The remaining duration of ( 2α)(m 2) symbols is allocated to unaided direct transmission from the source to the destination. 6

7 Again, we have < α /2 in this setting. In these protocols, the input-output relations are expressed as follows: y d = h sd x s + n d, y r = h sr x s2 + n r, y d2 = h sd x s2 + n d2, and y r d = h sd x s22 + h rd x r + n r d. (6) Above, x s,x s2,x s22, which have respective dimensions ( 2α)(m 2), α(m 2) and α(m 2), represent the source data vectors sent in direct transmission, cooperative transmission when relay is listening, and cooperative transmission when relay is transmitting, respectively. Note again that the source transmits all the time. x r is the relay s data vector with dimension α(m 2). y d,y d2,yd r are the corresponding received vectors at the destination, and y r is the received vector at the relay. The input vector x s now is defined as x s = [x T s,x T s2,x T s22] T and we again denote y d = [yd T yt d2 ]T. If we express the fading coefficients as h = ĥ + h in (6), we obtain the following input-output relations: y d = ĥsdx s + h sd x s + n d, y r = ĥsrx s2 + h sr x s2 + n r, y d2 = ĥsdx s2 + h sd x s2 + n d2, and (7) y r d = ĥsdx s22 + ĥrdx r + h sd x s22 + h rd x r + n r d. (8) IV. ACHIEVABLE RATES In this section, we provide achievable rate expressions for AF and DF relaying in both non-overlapped and overlapped transmission scenarios described in Section III. Achievable rate expressions are obtained by considering the estimate errors as additional sources of Gaussian noise. Since Gaussian noise is the worst uncorrelated additive noise for a Gaussian model [2], [2], achievable rates given in this section can be regarded as worst-case rates. We first consider AF relaying scheme. The capacity of the AF relay channel is the maximum mutual information between the transmitted signal x s and received signals y d and yd r given the estimates ĥsr, ĥsd, ĥrd: C = sup p xs ( ) m I(x s;y d,yd r ĥsr, ĥsd, ĥrd). (9) Note that this formulation presupposes that the destination has the knowledge of ĥsr. Hence, we assume that the value of ĥsr is forwarded reliably from the relay to the destination over low-rate control links. In general, solving the optimization problem in (9) and obtaining the channel capacity is a difficult task. Therefore, we concentrate on finding a lower bound on the capacity. A lower bound is obtained by replacing the product of the estimate error and the transmitted signal in the input-output relations with the worst-case noise with 7

8 the same correlation. In non-overlapped transmission, we consider z d = h sd x s + n d, z r = h sr x s2 + n r, z d2 = h sd x s2 + n d2, and z r d = h rd x r + n r d, (2) as the new noise vectors whose covariance matrices, respectively, are E{z d z d } = σ2 z d I = σ 2 hsd E{x s x s } + N I, E{z r z r } = σ2 z r I = σ 2 hsr E{x s2 x s2 } + N I, (2) E{z d2 z d2 } = σ2 z d2 I = σ 2 hsd E{x s2 x s2 } + N I, E{z r d zr d } = σ 2 z r d I = σ2 hrd E{x r x r } + N I. (22) Similarly, in overlapped transmission, we define z d = h sd x s + n d, z r = h sr x s2 + n r, z d2 = h sd x s2 + n d2, z r d = h sd x s22 + h rd x r + n r d, (23) as noise vectors with covariance matrices E{z d z d } = σ2 z d I = σ 2 hsd E{x s x s } + N I, E{z r z r } = σ2 z r I = σ 2 hsr E{x s2 x s } + N I, (24) E{z d2 z d2 } = σ2 z d2 I = σ 2 hsd E{x s2 x s2 }+N I, E{z r d zr d } = σ 2 z r d I = σ2 hsd E{x s22 x s22 }+σ2 hrd E{x r x r }+N I. An achievable rate expression is obtained by solving the following optimization problem which requires finding the worst-case noise: C I AF = inf sup p zd ( ),p zr ( ),p zd2 ( ),p z r( ) d p xs ( ) (25) m I(x s;y d,y r d ĥsr, ĥsd, ĥrd). (26) The following results provide I AF for both non-overlapped and overlapped transmission scenarios. Theorem : An achievable rate of AF in non-overlapped transmission scheme is given by ( ) ( [ ])] I AF = [( m E 2α)(m 2) log + P s ĥsd 2 + α(m 2) log + P s ĥsd 2 P + f s ĥsr 2, P r ĥrd 2 σz 2 d σz 2 d2 σz 2 r σz 2 d r (27) where f(x, y) = xy + x + y (28) P s ĥsd 2 σ 2 z d = P s ĥsd 2 σ 2 z d2 = δ s ( δ s )m 2 Ps 2σ4 sd /( α) ( δ s )mp s σsd 2 N /( α) + (m 2)(σsd 2 δ w sd 2 (29) smp s + N )N 8

9 P s ĥsr 2 σ 2 z r = δ s ( δ s )m 2 Ps 2σ4 sr /( α) w ( δ s )mp s σsrn 2 /( α) + (m 2)(σsrδ 2 sr 2 (3) s mp s + N )N P r ĥrd 2 σ 2 z r d = δ r ( δ r )m 2 P 2 r σ 4 rd /α ( δ r )mp r σ 2 rd N /α + (m 2)(σ 2 rd δ rmp r + N )N w rd 2. (3) In the above equations and henceforth, w sr CN(, ), w sd CN(, ), w rd CN(, ) denote independent, standard Gaussian random variables. Proof : Note that in non-overlapped AF relaying, I(x s ;y d,y r d ĥsr, ĥsd, ĥrd) = I(x s ;y d ĥsd) + I(x s2 ;y d2,y r d ĥsr, ĥsd, ĥrd) (32) where the first mutual expression on the right-hand side of (32) is for the direct transmission and the second is for the cooperative transmission. In the direct transmission, we have y d = ĥsdx s + z d. (33) In this setting, it is well-known that the worst-case noise z d is Gaussian [2] and x s with independent Gaussian components achieves inf sup p zd ( ) p xs ( ) I(x s ;y d ĥsd) = E [ ( 2α)(m 2) log ( + P s ĥsd 2 σ 2 z d )]. (34) Therefore, we now concentrate on the cooperative phase. For better illustration, we rewrite the channel input-output relationships in (4) and (5) for each symbol: y r [i] = ĥsrx s2 [i] + z r [i], y d2 [i] = ĥsdx s2 [i] + z d2 [i], (35) for i = + ( 2α)(m 2),..., ( α)(m 2), and y r d[i] = ĥrdx r [i] + z r d[i], (36) for i = ( α)(m 2)+,..., m 2. In AF, the signals received and transmitted by the relay have following relation: E[ x r x r [i] = βy r [i α(m 2)], where β 2 ] ĥsr 2 E[ x s2 2 ] + E[ z r 2 ]. (37) 9

10 Now, we can write the channel in the vector form y d2 [i] = ĥsd x s [i] + z r [i] z yd r [i + α(m 2)] ĥ rd βĥsr ĥ rd β d2 [i] }{{}}{{}}{{} zd r [i + α(m 2)] ˇy d [i] A B }{{} z[i] (38) for i = + ( 2α)(m 2),..., ( α)(m 2), With the above notation, we can write the input-output mutual information as I(x s2 ;y d2,y r d ĥsr, ĥsd, ĥrd) = ( α)(m 2) i=+( 2α)(m 2) I(x s [i]; ˇy d [i] ĥsr, ĥsd, ĥrd) = α(m 2)I(x s ; ˇy d ĥsr, ĥsd, ĥrd) (39) where in (39) we removed the dependence on i without loss of generality. Note that ˇy is defined in (38). Now, we can calculate the worst-case capacity by proving that Gaussian distribution for z r, z d2, and z r d provides the worst case. We employ techniques similar to that in [2]. Any set of particular distributions for z r, z d2, and z r d yields an upper bound on the worst case. Let us choose z r, z d2, and z r d complex Gaussian distributed. Then as in [6], to be zero mean inf sup p zr ( ),p zd2 ( ),p z r( ) d p xs2 ( ) I(x s ; ˇy d ĥsr, ĥsd, ĥrd) E log det ( I + (E( x s 2 )AA )(BE[zz ]B ) ) (4) where the expectation is with respect to the fading estimates. To obtain a lower bound, we compute the mutual information for the channel in (38) assuming that x s is a zero-mean complex Gaussian with variance E( x s 2 ), but the distributions of noise components z r, z d2, and z r d are arbitrary. In this case, we have I(x s ; ˇy d ; ĥsr, ĥsd, ĥrd) = h(x s ĥsr, ĥsd, ĥrd) h(x s ˇy d, ĥsr, ĥsd, ĥrd) log πee( x s 2 ) log πe var(x s ˇy d, ĥsr, ĥsd, ĥrd) (4) where the inequality is due to the fact that Gaussian distribution provides the largest entropy and hence h(x s ˇy d, ĥsr, ĥsd, ĥrd) log πe var(x s ˇy d, ĥsr, ĥsd, ĥrd). From [2], we know that var(x s ˇy d, ĥsr, ĥsd, ĥrd) E [ ] (x s ˆx s )(x s ˆx s ) ĥsr, ĥsd, ĥrd (42) for any estimate ˆx s given ˇy d, ĥsr, ĥsd, and ĥrd. If we substitute the LMMSE estimate ˆx s = R xy R y ˇy d into

11 (4) and (42), we obtain 3 I(x s ; ˇy d ĥsr, ĥsd, ĥrd) E log det ( I + (E[ x s 2 ]AA )(BE[zz ]B ) ). (43) Since the lower bound (43) applies for any noise distribution, we can easily see that inf sup p zr ( ),p zd2 ( ),p z r( ) d p xs2 ( ) From (4) and (44), we conclude that I(x s ; ˇy d ĥsr, ĥsd, ĥrd) E log det ( I + (E[ x s 2 ]AA )(BE[zz ]B ) ). (44) inf sup p zr ( ),p zd2 ( ),p z r( ) d p xs2 ( ) In (46), P s and P r σ 2 z d2, σ 2 z r, σ 2 z r d I(x s ; ˇy d ĥsr, ĥsd, ĥrd) = E log det ( I + (E[ x s 2 ]AA )(BE[zz ]B ) ) (45) = E [ log ( + P s ĥsd 2 σ 2 z d2 + f [ ])] P s ĥsr 2, P r ĥrd 2. (46) σz 2 r σz 2 d r are the powers of source and relay symbols and are given in (9) and (). Moreover, are the variances of the noise components defined in (2). Now, combining (26), (32), (34), and (46), we obtain the achievable rate expression in (27). Note that (29) (3) are obtained by using the expressions for the channel estimates in (5) (7) and noise variances in (2) and (22). Theorem 2: An achievable rate of AF in overlapped transmission scheme is given by [ ( I AF = m E ( 2α)(m 2) log( + P s2 ĥsd 2 )+(m 2)α log + P s2 ĥsd 2 σz 2 d σz 2 d2 ( P + q s2 ĥsd 2 where q(.) is defined as q(a, b, c, d) = (+a)b(+c). Moreover +c+d P s2 ĥsd 2 σ 2 z d = P s2 ĥsd 2 σ 2 z d2 = P s2 ĥsr 2 σ 2 z r =, P s2 ĥsd 2, P s2 ĥsr 2, P r ĥrd 2 σz 2 d2 σz 2 σ d r z 2 r σz 2 d r ( P + f s2 ĥsr 2, P r ĥrd 2 σz 2 r σ 2 z r d ) ) )] (47) δ s ( δ s )m 2 P 2 s σ4 sd ( δ s )mp s σ 2 sd N + (m 2)(σ 2 sd δ smp s + N )N w sd 2 (48) δ s ( δ s )m 2 P 2 s σ 4 sr ( δ s )mp s σ 2 sr N + (m 2)(σ 2 sr δ smp s + N )N w sr 2 (49) P s2 ĥsd 2 σ 2 z r d = δ s ( δ s )m 2 P 2 s σ 4 sd (σ2 rd δ rmp r + N ) w sd 2 (m 2)(σ 2 sd δ smp s +N )(σ 2 rd δ rmp r +N )N + ( δ r )mp r σ 2 rd N (σ 2 sd δ smp s + N )/α + ( δ s )mp s σ 2 sd N (σ 2 rd δ rmp r + N ) (5) 3 Here, we use the property that det(i + AB) = det(i + BA).

12 P r ĥrd 2 σ 2 z r d = δ r ( δ r )m 2 P 2 r σ4 rd (σ2 sd δ smp s + N )/α w rd 2 (m 2)(σ 2 sd δ smp s +N )(σ 2 rd δ rmp r +N )N + ( δ r )mp r σ 2 rd N (σ 2 sd δ smp s + N )/α + ( δ s )mp s σ 2 sd N (σ 2 rd δ rmp r + N ) (5) Proof : Note that the only difference between the overlapped and non-overlapped transmissions is that source continues its transmission as the relay transmits. As a result, the power of each source symbol is now P s2 given in (9). Additionally, when both the source and relay are transmitting, the received signal at the destination is yd r = ĥsdx s22 + ĥrdx r + h sd x s22 + h rd x r +n r d. The input-output mutual information in one block is I(x s ;y d,y r d ĥsr, ĥsd, ĥrd) = I(x s ;y d ĥsd) + I(x s2,x s22 ;y d2,y r d ĥsr, ĥsd, ĥrd). (52) The first term on the right-hand-side of (52) corresponds to the mutual information of the direct transmission and is the same as that in non-overlapped transmission. Hence, the worst-case rate expression obtained in the proof of Theorem is valid for this case as well. In the cooperative phase, the input-output relation for each symbol can be written in the following matrix form: y d2 [i] = ĥsd x s [i] + z r [i] z yd r [i + α(m 2)] ĥ rd βĥsr ĥ sd x s [i + α(m 2)] ĥ rd β d2 [i] }{{}}{{}}{{}}{{} zd r [i + α(m 2)] ˇy d [i] A ˇx s[i] B }{{} z[i] where i = +( 2α)(m 2),..., ( α)(m 2) and β E[ x r 2 ] ĥsr 2 E[ x s 2 ]+E[ z r 2 ] (53). Note that we have defined x s = [x T s,xt s2,xt s22 ]T, and the expression in (53) uses the property that x 2 (j) = x s (j + ( 2α)(m 2)) and x s22 (j) = x s (j + ( α)(m 2)) for j =,...,α(m 2). The input-output mutual information in the cooperative phase can now be expressed as I(x s2,x s22 ;y d2,y r d ĥsr, ĥsd, ĥrd) = ( α)(m 2) i=+( 2α)(m 2) I(ˇx s [i]; ˇy d [i] ĥsr, ĥsd, ĥrd) = α(m 2)I(ˇx s ; ˇy d ĥsrĥsd, ĥrd) (54) where in (54) we removed the dependence on i without loss of generality. Note that ˇx and ˇy are defined in (53). As shown in proof of Theorem, the worst-case achievable rate for cooperative transmission is inf sup p zr ( ),p zd2 ( ),p z r( ) d p xs2 ( ) I(ˇx s ; ˇy d ĥsr, ĥsd, ĥrd) = E log det ( I + (E[ˇx sˇx s ]AA )(BE[zz ]B ) ). (55) 2

13 Using the definitions in (53) and evaluating the log det expression in (55), and combining the direct transmission worst-case achievable rate, we arrive to (47). (48) (5) are obtained by using the expressions for the channel estimates in (5) (7) and noise variances in (24) and (25). Next, we consider DF relaying scheme. In DF, there are two different coding approaches [7], namely repetition coding and parallel channel coding. We first consider repetition channel coding scheme. The following results provide achievable rate expressions in both non-overlapped and overlapped transmission scenarios. Theorem 3: An achievable rate expression for DF with repetition channel coding for non-overlapped transmission scheme is given by I DFr = ( 2α)(m 2) E m [ log ( + P s ĥsd 2 σ 2 z d )] + α(m 2) m min{i, I 2 } (56) where [ I = E log ( + P s ĥsr 2)] [, and I σz 2 2 = E log ( + P s ĥsd 2 + P r ĥrd 2 ) ]. (57) r σz 2 d2 σz 2 d r Moreover, P s ĥsd 2, P s ĥsr 2, and P r ĥrd 2 are the same as defined in (29) (3). σzr 2 σ 2 z d r Proof : For DF with repetition coding in non-overlapped transmission, an achievable rate expression is, P σz 2 s ĥsd 2 σ 2 d z d2 { } I(x s ;y d ĥsd) + min I(x s2 ;y r ĥsr), I(x s2 ;y d,yd r ĥsd, ĥrd). (58) Note that the first and second mutual information expressions in (58) are for the direct transmission between the source and destination, and direct transmission between the source and relay, respectively. Therefore, as in the proof of Theorem, the worst-case achievable rates can be immediately seen to be equal to the first term on the right-hand side of (56) and I, respectively. In repetition coding, after successfully decoding the source information, the relay transmits the same codeword as the source. As a result, the input-output relation in the cooperative phase can be expressed as where β y d [i] = ĥsd z x s [i] + d2 [i]. (59) yd r [i + α(m 2)] ĥ rd β zd r [i + α(m 2)] }{{}}{{}}{{} ˇy d [i] A z[i] E[ xr 2 ] E[ x s 2 ]. From (59), it is clear that the knowledge of ĥsr is not required at the destination. We can easily see that (59) is a simpler expression than (38) in the AF case, therefore we can adopt the same 3

14 methods as employed in the proof of Theorem to show that Gaussian noise is the worst noise and I 2 is the worst-case rate. Theorem 4: An achievable rate expression for DF with repetition channel coding for overlapped transmission scheme is given by where I = E [ log ( I DFr = + P s2 ĥsr 2 σ 2 z r ( 2α)(m 2) E m )], I 2 = E [ [ log log ( ( + P s2 ĥsd 2 σ 2 z d )] + (m 2)α m min{i, I 2 } (6) )] + P s2 ĥsd 2 + P r ĥrd 2 + P s2 ĥsd 2 + P s2 ĥsd 2 P s2 ĥsd 2. σz 2 d2 σz 2 σ 2 d r z σ d r z 2 d2 σz 2 d r (6) P s2 ĥsd 2, P σz 2 s2 ĥsd 2 σ 2 d z d2, P s2 ĥsr 2, P σzr 2 s2 ĥsd 2, P σ 2 r ĥrd 2 have the same expressions as in (48) (5). z d r σ 2 z d r Proof : Note that in overlapped transmission, source transmits over the entire duration of (m 2) symbols, and hence the channel input-output relation in the cooperative phase is expressed as follows: y d [i] = ĥsd x s [i] z + d22 [i]. (62) yd r [i + α(m 2)] ĥ rd β }{{}}{{ ĥsd x s [i + α(m 2)] zd2 r [i + α(m 2)] }}{{}}{{} ˇy d [i] A ˇx s[i] z[i] The result is immediately obtained using the same techniques as in the proof of Theorem 2. Finally, we consider DF with parallel channel coding and assume that non-overlapped transmission scheme is adopted. From [], we note that an achievable rate expression is min{( α)i(x s ;y r ĥsr), ( α)i(x s ;y d ĥsd) + αi(x r ;y r d ĥrd)}. Note that we do not have separate direct transmission in this relaying scheme. Using similar methods as before, we obtain the following result. The proof is omitted to avoid repetition. Theorem 5: An achievable rate of non-overlapped DF with parallel channel coding scheme is given by { [ ( )] [ ( )] ( α)(m 2) I DFp = min E log + P s ĥsr 2 ( α)(m 2), E log + P s ĥsd 2 + m σz 2 r m σz 2 d2 [ ( )]} α(m 2) m E log + P r ĥrd 2 (63) σz 2 d r 4

15 where P s ĥsd 2, P σz 2 s ĥsr 2, and P r ĥrd 2 σ 2 d2 zr σ 2 z d r are given in (29)-(3). V. OPTIMAL RESOURCE ALLOCATION Having obtained achievable rate expressions in Section IV, we now identify optimal resource allocation strategies that maximize the rates. We consider three resource allocation problems: ) power allocation between the training and data symbols; 2) time/bandwidth allocation to the relay; 3) power allocation between the source and relay under a total power constraint. We first study how much power should be allocated for channel training. In nonoverlapped AF, it can be seen that δ r appears only in P r ĥrd 2 σ 2 z r d in the achievable rate expression (27). Since f(x, y) = xy +x+y is a monotonically increasing function of y for fixed x, (27) is maximized by maximizing P r ĥrd 2. We can maximize P r ĥrd 2 σ 2 z d r is given below: by maximizing the coefficient of the random variable w rd 2 in (3), and the optimal δ r σ 2 z r d δ opt r = mprσ2 rd αmn + 2αN + p α(m 2)(m 2 P rσrd 2 αn + m2 Pr 2 σrd 4 + αmn2 + mprσ2 rd N 2mPrσ2 rdαn 2Nα) mp rσrd 2 ( + αm 2α). (64) Optimizing δ s is more complicated as it is related to all the terms in (27), and hence obtaining an analytical solution is unlikely. A suboptimal solution is to maximize P s ĥsd 2 and P σz 2 s ĥsr 2 separately, and obtain two σ 2 d zr solutions δ subopt s, and δ subopt s,2, respectively. Note that expressions for δ subopt s, and δ subopt s,2 are exactly the same as that in (64) with P r, α, and σ rd replaced by P s, ( α), and and σ sr, respectively. When the source-relay channel is better than the source-destination channel and the fraction of time over which direct transmission is performed is small, P s ĥsr 2 is a more dominant factor and δ subopt σzr 2 s,2 is a good choice for training power allocation. Otherwise, δ subopt s, might be preferred. Note that in non-overlapped DF with repetition and parallel coding, P r ĥrd 2 is the only term that includes δ σ 2 r. Therefore, similar results and discussions apply. For z d r instance, the optimal δ r has the same expression as that in (64). Figure plots the optimal δ r as a function of σ rd for different relay power constraints P r when m = 5 and α =.5. It is observed in all cases that the allocated training power monotonically decreases with improving channel quality and converges to α(m 2) αm 2α.69 which is independent of P r. In overlapped transmission schemes, both δ s and δ r appear in more than one term in the achievable rate expressions. Therefore, we resort to numerical results to identify the optimal values. Figures 2 and 3 plot the achievable rates as a function of δ s and δ r for overlapped AF. In both figures, we have assumed that 5

16 =, σ sr = 2, σ rd = and m = 5, N =, α =.5. While Fig. 2, where P s = 5 and P r = 5, considers high SNRs, we assume that P s =.5 and P r =.5 in Fig. 3. In Fig. 2, we observe that increasing δ s will increase achievable rate until δ s.. Further increase in δ s decreases the achievable rates. On the other hand, rates always increase with increasing δ r. This indicates that cooperation is not beneficial in terms of achievable rates and direct transmission should be preferred. On the other hand, in the low-power regime considered in Fig. 3, the optimal values of δ s and δ r are approximately.8 and.32, respectively. Hence, the relay in this case helps to improve the rates. Next, we analyze the effect of the degree of cooperation on the performance in AF and repetition DF. Figures 4-7 plot the achievable rates as a function of α which gives the fraction of total time/bandwidth allocated to the relay. Achievable rates are obtained for different channel qualities given by the standard deviations, σ sr, and σ rd of the fading coefficients. We observe that if the input power is high, α should be either.5 or close to zero depending on the channel qualities. On the other hand, α =.5 always gives us the best performance at low SNR levels regardless of the channel qualities. Hence, while cooperation is beneficial in the low-snr regime, noncooperative transmissions might be optimal at high SNRs. We note from Fig. 4 that cooperation starts being useful as the source-relay channel variance σsr 2 increases. Similar results are also observed in Fig 5. Hence, the source-relay channel quality is one of the key factors in determining the usefulness of cooperation in the high SNR regime. In Fig. 8, we plot the achievable rates of DF parallel channel coding, derived in Theorem 5. We can see from the figure that the best performance is obtained when the source-relay channel quality is high (i.e., when =, σ sr =, σ rd = 2). Additionally, we observe that as the source-relay channel improves, more resources need to be allocated to the relay to achieve the best performance. We note that significant improvements with respect to direct transmission (i.e., the case in which α ) are obtained. Finally, we can see that when compared to AF and DF with repetition coding, DF with parallel channel coding achieves higher rates. On the other hand, AF and repetition coding DF have advantages in the implementation. Obviously, the relay, which amplifies and forwards, has a simpler task than that which decodes and forwards. Moreover, as pointed out in [4], if AF or repetition coding DF is employed in the system, the architecture of the destination node is simplified because the data arriving from the source and relay can be combined rather than stored separately. In certain cases, source and relay are subject to a total power constraint. Here, we introduce the power allocation coefficient θ, and total power constraint P. P s and P r have the following relations: P s = θp, 6

17 P r = ( θ)p, and P s +P r P. Next, we investigate how different values of θ, and hence different power allocation strategies, affect the achievable rates. An analytical results for θ that maximizes the achievable rates is difficult to obtain. Therefore, we again resort to numerical analysis. In all numerical results, we assume that α =.5 which provides the maximum of degree of cooperation. First, we consider the AF. The fixed parameters we choose are P =, N =, δ s =., δ r =.. Fig. 9 plots the achievable rates in the overlapped transmission scenario as a function of θ for different channel conditions, i.e., different values of σ sr, σ rd, and. We observe that the best performance is achieved as θ. Hence, even in the overlapped scenario, all the power should be allocated to the source and direct transmission should be preferred at these high SNR levels. Note that if direct transmission is performed, there is no need to learn the relay-destination channel. Since the time allocated to the training for this channel should be allocated to data transmission, the real rate of direct transmission is slightly higher than the point that the cooperative rates converge as θ. For this reason, we also provide the direct transmission rate separately in Fig. 9. Further numerical analysis has indicated that direct transmission over performs non-overlapped AF, overlapped and non-overlapped DF with repetition coding as well at this level of input power. On the other hand, in Fig. which plots the achievable rates of non-overlapped DF with parallel coding as a function of θ, we observe that direct transmission rate, which is the same as that given in Fig. 9, is exceeded if σ sr = and hence the source-relay channel is very strong. The best performance is achieved when θ.7 and therefore 7% of the power is allocated to the source. Figs., 2, and 3 plot the non-overlapped achievable rates when P =. In all cases, we observe that performance levels higher than that of direct transmission are achieved unless the qualities of the source-relay and relay-destination channels are comparable to that of the source-destination channel (e.g., =, σ sr = 2, σ rd = ). Moreover, we note that the best performances are attained when the sourcerelay and relay-destination channels are both considerably better than the source-destination channel (i.e., when =, σ sr = 4, σ rd = 4). As expected, highest gains are obtained with parallel coding DF although repetition coding incur only small losses. Finally, Fig. 4 plot the achievable rates of overlapped AF when P =. Similar conclusions apply also here. However, it is interesting to note that overlapped AF rates are smaller than those achieved by non-overlapped AF. This behavior is also observed when DF with repetition coding is considered. Note that in non-overlapped transmission, source transmits in a shorter duration of time with higher power. This signaling scheme provides better performance as expected because it is well-known that flash signaling achieves the capacity in the low-snr regime in imperfectly known channels [8]. 7

18 VI. ENERGY EFFICIENCY Our analysis has shown that cooperative relaying is generally beneficial in the low-power regime, resulting in higher achievable rates when compared to direct transmission. In this section, we provide an energy efficiency perspective and remark that care should also be taken when operating at very low SNR values. The least amount of energy required to send one information bit reliably is given by 4 E b N = SNR C(SNR) where C(SNR) is the channel capacity in bits/symbol. In our setting, the capacity will be replaced by the achievable rate expressions and hence the resulting bit energy, denoted by E b,u N, provides the least amount of normalized bit energy values in the worst-case scenario and also serves as an upper bound on the achievable bit energy levels in the channel. We note that in finding the bit energy values, we assume that SNR = P/N where P = P r + P s is the total power. The next result provides the asymptotic behavior of the bit energy as SNR decreases to zero. Theorem 6: The normalized bit energy in all relaying schemes grows without bound as the signal-to-noise ratio decreases to zero, i.e., E b,u N = lim I= SNR SNR I(SNR) = I() =. (65) Proof : The key point to prove this theorem is to show that when SNR, the mutual information decreases as SNR 2, and hence I() =. This can be easily shown because when P, in all the terms P s ĥsd 2 σ 2 z d, P s ĥsd 2 σ 2 z d2, P s ĥsr 2, P σzr 2 r ĥrd 2, P σ 2 s2 ĥsd 2, P z d r σz 2 s2 ĥsr 2, P σ 2 s2 ĥsd 2, and P d2 zr σ 2 r ĥrd 2 in Theorems -5, the denominator goes to a z d r σ 2 z d r constant while the numerator decreases as P 2. Hence, these terms diminish as SNR 2. Since log( + x) = σ 2 z d, P s2 ĥsd 2 x+o(x) for small x, we conclude that the achievable rate expressions also decrease as SNR 2 as SNR vanishes. Theorem 6 indicates that it is extremely energy-inefficient to operate at very low SNR values. We identify the most energy-efficient operating points in numerical results. We choose the following numerical values for the fixed parameters: δ s = δ r =., =, σ sr = 4, σ rd = 4, α =.5, and θ =.6. Fig. 5 plots the bit energy curves as a function of SNR for different values of m in the non-overlapped AF case. We can see from the figure that the minimum bit energy, which is achieved at a nonzero value of SNR, decreases with increasing m and is achieved at a lower SNR value. Fig. 6 shows the minimum bit energy for different relaying schemes with overlapped or non-overlapped transmission techniques. We observe that 4 Note that E b N is the bit energy normalized by the noise power spectral level N. 8

19 the minimum bit energy decreases with increasing m in all cases. We realize that DF is in general much more energy-efficient than AF. Moreover, we note that employing non-overlapped rather than overlapped transmission improves the energy efficiency. We further remark that the performances of non-overlapped DF with repetition coding and parallel coding are very close. VII. CONCLUSION In this paper, we have studied the imperfectly-known fading relay channels. We have assumed that the source-destination, source-relay, and relay-destination channels are not known by the corresponding receivers a priori, and transmission starts with the training phase in which the channel fading coefficients are learned with the assistance of pilot symbols, albeit imperfectly. Hence, in this setting, relaying increases the channel uncertainty in the system, and there is increased estimation cost associated with cooperation. We have investigated the performance of relaying by obtaining achievable rates for AF and DF relaying schemes. We have considered both non-overlapped and overlapped transmission scenarios. We have controlled the degree of cooperation by varying the parameter α. We have identified the optimal resource allocation strategies using the achievable rate expressions. We have observed that if the source-relay channel quality is low, then cooperation is not beneficial and direct transmission should be preferred at high SNRs. On the other hand, we have seen that relaying generally improves the performance at low SNRs. We have noted that DF with parallel coding provides the highest rates. Additionally, under total power constraints, we have identified the optimal power allocation between the source and relay. We have again pointed out that relaying degrades the performance at high SNRs unless DF with parallel channel coding is used and the source-relay channel quality is high. The benefits of relaying is again demonstrated at low SNRs. We have noted that non-overlapped transmission is superior compared to overlapped one in this regime. Finally, we have considered the energy efficiency in the low-power regime, and proved that the bit energy increases without bound as SNR diminishes. Hence, operation at very low SNR levels should be avoided. From the energy efficiency perspective, we have again observed that non-overlapped transmission provides better performance than overlapped transmission. We have also noted that DF is more energy efficient than AF. REFERENCES [] E. C. van der Meulen, Three-terminal communication channels, Adv. Appl. Probab., vol. 3, pp. 2-54, 97. [2] T. M. Cover and A.A. El Gamal, Capacity theorems for the relay channel, IEEE Trans. Inf. Theory, vol. IT-25, no. 5, pp , Sep

20 [3] A. A. El Gamal and M. Aref, The capacity of the semideterministic relay channel, IEEE Trans. Inf. Theory, vol. IT-28, no. 3, pp , May 982. [4] A. Sendonaris, E. Erkip, and B. Aazhang, User cooperation diversity-part I: System description, IEEE Trans. Commun., vol. 5, no., pp , Nov. 23. [5] A. Sendonaris, E. Erkip, and B. Aazhang, User cooperation diversity-part II: Implementation aspects and performance analysis, IEEE Trans. Commun., vol. 5, no., pp , Nov. 23. [6] J.N. Laneman, D.N.C. Tse, G.W. Wornel Cooperative diversity in wireless networks: Efficient protocols and outage behavior, IEEE Trans. Inform. Theory, vol.5, pp , Dec. 24 [7] J. N. Laneman, Cooperation in wireless networks: Principles and applications, Chapter, Cooperative Diversity: Models, Algorithms, and Architectures, Springer, 26. [8] R.U. Nabar, H. Bolcskei, F.W. Kneubuhler, Fading Relay Channels:Performance Limits and Space-Time Signal Design, IEEE J.Select. Areas Commun vol.22, no.6 pp. 99-9, Aug. 24 [9] A. Host-Madsen, Capacity bounds for cooperative diversity, IEEE Trans. Inform. Theory, vol. 52, no. 4, pp , Apr. 26. [] G. Kramer, M. Gastpar, and P. Gupta, Cooperative strategies and capacity theorems for relay networks, IEEE Trans. Inform. Theory, vol. 5, no. 9, pp , Sept. 25. [] Y. Liang and V. V. Veeravalli, Gaussian orthogonal relay channels: Optimal resource allocation and capacity, IEEE Trans. Inform. Theory, vol. 5, no. 9, pp , Sept. 25. [2] P. Mitran, H. Ochiai, and V. Tarokh, Space-time diversity enhancements using collaborative communications, IEEE Trans. Inform. Theory, vol. 5, no. 6, pp , June 25. [3] Y. Yao, X. Cai, and G. B. Giannakis, On energy efficiency and optimum resource allocation of relay tranmissions in the low-power regime, IEEE Trans. Wireless Commun., vol. 4, no. 6, pp , Nov. 25. [4] G. Kramer, I. Marić, and R. D. Yates, Cooperative communications, Foundations and Trends in Networking. Hanover, MA: NOW Publishers Inc., vol., no. 3-4, pp , 26. [5] Special Issue on Models, Theory, and Codes for Relaying and Cooperation in Communication Networks, IEEE Tran. Inform. Theory, vol.53, no.9, Oct. 27. [6] B.Wang, J.Zhang, L.Zheng, Achievable rates and scaling laws of power-constrained wireless sensory relay networks IEEE Tran. Inform. Theory, vol.52, no.9, Sep.26. [7] A.S. Avestimehr, D.N.C. Tse, Outage Capacity of the Fading Relay Channel in the Low-SNR Regime, IEEE Trans. Inform. Theory, vol.53, pp. 4-45, April 27. [8] S. Verdú, Spectral efficiency in the wideband regime, IEEE Trans. Inform. Theory, vol. 48, pp , June 22. [9] M. Medard The effect upon channel capacity in wireless communication of perfect and imperfect knowledge of the channel, IEEE Trans. Inform. Theory, vol.46, pp , May 2. [2] B. Hassibi, B. M. Hochwald, How much training is needed in multiple-antenna wireless link?, IEEE Trans. Inform. Theory, vol.49, pp , April 23, [2] L. Tong, B. M. Sadler, and M. Dong, Pilot-assisted wireless transmission, IEEE Signal Processing Mag., pp. 2-25, Nov. 24. [22] J. Zhang, M.C. Gursoy, To cooperate, or not to cooperate in imperfectly known fading channels, submitted to the 9th IEEE International Workshop on Signal Processing Advances in Wireless Communications (SPAWC), 28. [23] S. Akin, M.C. Gursoy, Achievable rates and training optimization for fading relay channels with memory, Proc. of the 42nd Annual Conference on Information Sciences and Systems (CISS), Princeton University, 28. [24] J. Zhang, M.C. Gursoy, Achievable rates and optimal resource allocation for imperfectly-known relay channels, Proc. of the 45th Annual Allerton Conference on Communication, Control and Computing, Sept. 27. [25] M.C. Gursoy, An energy efficiency perspective on training for fading channels, Proceedings of the IEEE ISIT 27. 2

21 Pr=. Pr= Pr=5 Pr=2 Pr=2.35 δ r σ rd Fig.. δ r vs. σ rd for different values of P r when m = Achievable Rates (bits/symbol) δ s δ r Fig. 2. Overlapped AF achievable rates vs. δ s and δ r when P s = P r = 5 2

22 .4 Achievable Rates (bits/symbol) δ s.2.4 δ r.6.8 Fig. 3. Overlapped AF achievable rates vs. δ s and δ r when P s = P r =.5 Achievable Rates (bits/symbol) = σ rd =2 =6 σ rd =3 =4 σ rd =4 =2 σ rd = α Fig. 4. Overlapped AF achievable rate vs. α when P s = P r = 5, δ s = δ r =., m = 5. 22

23 Achievable Rates (bits/symbol) = σ rd =2 =6 σ rd =3 3.2 σ = σ =4 σ =4 sd sr rd σ = σ =2 σ = sd sr rd α Fig. 5. Overlapped DF with repetition coding achievable rate vs. α when P s = P r = 5, δ s = δ r =., m = 5. Achievable Rates (bits/symbol) = σ rd =2 =6 σ rd =3 =4 σ rd =4 =2 σ rd = α Fig. 6. Overlapped AF achievable rate vs. α when P s = P r =.5, δ s = δ r =., m = 5. 23

24 Achievable Rates (bits/symbol) = σ rd =2 =6 σ rd =3 =4 σ rd =4 σ = σ =2 σ = sd sr rd α Fig. 7. Overlapped DF with repetition coding achievable rate vs. α when P s = P r =.5, δ s = δ r =., m = 5. Achievable Rates (bits/symbol) σ = σ = σ =2 sd sr rd =6 σ rd =3 =4 σ rd =4 σ = σ =2 σ = sd sr rd α Fig. 8. Non-overlapped DF parallel coding achievable rate vs. α when P s = P r =.5, δ s = δ r =., m = 5. 24

25 6 5 Achievable Rates (bits/symbol) σ =,σ =, σ =2 sd sr rd σ =,σ =6, σ =3 sd sr rd σ sd =4, σ rd =4 =2, σ rd = Real Rate of Direct Transmission θ Fig. 9. Overlapped AF achievable rate vs. θ. P =, m = Achievable Rates (bits/symbol) =, σ rd =2 =6, σ rd =3 =4, σ rd =4 =2, σ rd = θ Fig.. Non-overlapped Parallel coding DF rate vs. θ. P =, m = 5. 25

26 .9.8 Achievable Rates (bits/symbol) σ.3 sd =, σ rd =2 =6, σ rd =3.2 σ =,σ =4, σ =4 sd sr rd. σ =,σ =2, σ = sd sr rd Real Rate of Direct Transmission θ Fig.. Non-overlapped AF achievable rate vs. θ. P =, m = 5..5 Achievable Rate (bits/symbol).5 =, σ rd =2 σ =,σ =6, σ =3 sd sr rd σ =,σ =4, σ =4 sd sr rd =2, σ rd = θ Fig. 2. Non-overlapped Repetition coding DF rate vs. θ. P =, m = 5. 26

27 .6.4 Achievable Rate (bits/symbol) σ =,σ =, σ =2 sd sr rd σ =,σ =6, σ =3 sd sr rd.2 =4, σ rd =4 =2, σ rd = θ Fig. 3. Non-overlapped Parallel coding DF rate vs. θ. P =, m = Achievable Rates (bits/symbol) =, σ rd =2 =6, σ rd =3 =4, σ rd =4 =2, σ rd = θ Fig. 4. Overlapped AF achievable rate vs. θ. P =, m = 5. 27