Introduction and Background

Analog CMOS Integrated Circuit Design Introduction and Background Dr. Jawdat Abu-Taha Department of Electrical and Computer Engineering Islamic University of Gaza jtaha@iugaza.edu.ps 1

Marking Assignments 10% (4 to 6) Midterm 25% Project 20% Final Exam 45% 2

References Recommended Textbook: Behzad Razavi, Design of Analog CMOS Integrated Hill, 2001 Some other useful references: Circuits, McGraw- T. Chan Carusone, D. Johns and K. Martin, Analog Integrated Circuit Design, 2 nd Edition, John Wiley, 2011 P. Gray, P. Hurst, S. Lewis, and R. Meyer, Analysis and Design of Analog Integrated Circuits, Edition, John Wiley, 2009 SM 5 th D. Holberg and P. Allen, CMOS Analog Circuit Design, Oxford University Press, 2011 R. Jacob Baker, CMOS Circuit Design, Layout, and Simulation, Edition, Wiley-IEEE Press, 2010 A. Sedra and K.C. Smith, Microelectronic Circuits, Oxford University Press, 2004, 2009 Journal Circuits 3 rd 5 th or 6 th Edition, 3 rd Edition, and conference articles including IEEE Journal of Solid-State and International Solid-State Circuits Conference 3

Why Analog? Most of the physical signals are analog in nature! Although digital is great we need an analog interface to convert physical signals from analog to digital Also, in some application after processing the signals in digital we need to convert them back to analog. Thus in many applications analog and mixed-signal circuits performance bottlenecks. domain, are the Also with constant process improvements the boundary of between high-speed digital and analog circuits becomes more and more fuzzy! That is why analog and mixed-signal designers are still and hopefully will be in demand for the foreseeable future. SM 5

Data Converter Typical Real World System AFE DSP Example: G Filter ADC DSP SM 6

Intel 45 nm Process http://blog.oregonlive.com/siliconforest/2007/11/intel11.pdf SM 8

1. 2. 3. Suggested Reading Structure of MOS Transistors Threshold Voltage Background 4. Long-Channel Current Equations 5. Regions of Operation 6. Transconductance 7. Second-Order Effects 8. Short-Channel Effects 9. MOS Layout 10.Device Capacitances 11.Small-signal Models 12.Circuit Impedance 13.Equivalent Transconductance SM EECE 488 Set 1: Introduction and Background 9

Suggested Reading Most of the material in this set are based on Chapters 2, 16, and 17 of the Razavi s CMOS Integrated Circuits book: Design of Analog Many of the figures in this set are from Design of Analog CMOS Integrated Circuits, McGraw-Hill, 2001, unless otherwise noted. SM 10

Transistor Transistor stands for Transistor are semiconductor devices that can be classified as Bipolar Junction Transistors (BJTs) Field Effect Transistors (FETs) Depletion-Mode FETs or (e.g., JFETs) Enhancement-Mode FETs (e.g., MOSFETs) 11

Simplistic Model MOS transistors have three terminals: Gate, Source, and Drain The voltage of the Gate terminal determines between Source and Drain (Short or Open). Thus, MOS devices behave like a switch the type of connection NMOS PMOS V G high V G low Device is ON D is shorted to S Device is OFF D & S are disconnected Device is OFF D & S are disconnected Device is ON D is shorted to S 12

Physical Structure - 1 Source and Drain terminals are identical charge carriers, and Drain receives them. MOS devices have in fact 4 terminals: Source, Drain, Gate, Substrate (bulk) except that Source provides 13

Physical Structure - 2 Charge Carriers are electrons in NMOS devices, PMOS devices. Electrons have a higher mobility than holes So, NMOS devices are faster than PMOS devices We rather to have a p-type substrate?! and holes in L D : Due to Side Diffusion Poly-silicon used instead of Metal for fabrication reasons Actual length of the channel (L eff ) is less than the length of gate 14

Physical Structure - 3 N-wells allow both NMOS and PMOS devices to reside on the same piece of die. As mentioned, NMOS Source, Drain, Gate, and PMOS devices have Substrate (bulk) 4 terminals: In order to have all PN junctions reverse-biased, substrate of NMOS is connected to the most negative voltage, and substrate of PMOS is connected to the most positive voltage. 15

Physical Structure - 4 MOS transistor Symbols: In NMOS Devices: Source e lect ron Drain Current flows from Drain to Source In PMOS Devices: Source h ole Drain Current flows from Source to Drain Current flow determines which terminal is Source and which one is Drain. Equivalently, source and drain can be determined based on their relative voltages. 16

Threshold Voltage - 1 Consider an NMOS: as the gate voltage is increased, the surface under the gate is depleted. If the gate voltage increases more, free electrons appear under the gate and a conductive channel is formed. (a) An NMOS driven by a gate voltage, (b) formation of depletion region, (c) onset of inversion, and (d) channel formation As mentioned channel under before, in NMOS devices charge carriers in the the gate are electrons. 17

Threshold Voltage - 2 Intuitively, the threshold voltage is the gate voltage that forces the interface (surface under the gate) to be completely depleted of charge (in NMOS the interface is as much n-type as the substrate is p-type) Increasing induces an gate voltage above this threshold (denoted by V TH or V t ) inversion layer (conductive channel) under the gate. 18

Threshold Voltage - 3 Analytically: V TH MS 2 F Q dep C ox Where: MS Built - inpotential gate Silicon the difference between the work functions of the polysilico n gate and the silicon substrate Work Function (electrostatic potential) F K T q N ln sub ni Charge in the depletion region Q dep si F sub 4 q N 19

Threshold Voltage - 4 In practice, the native threshold value may not be suited for circuit design, e.g., V TH may be zero and the device may be on for any positive gate voltage. Typically threshold voltage is adjusted by ion implantation into the channel surface (doping P-type material will increase V TH of NMOS devices). When V DS is zero, there is no horizontal electric field present in the channel, and therefore no current between the source to the drain. When V DS is more than zero, there is some horizontal electric field which causes a flow of electrons from source to drain. 20

Long Channel Current Equations - 1 The voltage of the surface under the gate, V(x), depends on the voltages of Source and Drain. If V DS is zero, V D = V S =V(x). The charge density Q d (unit C/m) is uniform. Q C V C oxwlv GS V TH Q d L L L Q d WC ox (V GS V TH ) If V DS is not zero, the channel is tapered, and V(x) is not constant. The charge density depends on x. Q d (x) WC ox (V GS V (x) V TH ) 21

Current : I Long Channel Current Equations - 3 dq dq dx Q d velocity dt dx dt Velocity in terms of V(x): velocity E, E velocity ( dv (x) ) dx Q d in terms of V(x): dv dt Q d (x) WC ox (V GS V (x) V TH ) Current in terms of V(x): I WC [V V(x) V ] D ox GS TH n dv (x) dx L V DS I D dx WC ox n [V GS V ( x) V TH ]dv x0 V 0 Long-channel current equation: W 1 V 2 I D n C ox [(V GS V TH )V DS DS ] L 2 Microelectronic Circuits, 2004 Oxford University Press 22

Long Channel Current Equations - 4 If V DS V GS -V TH we say the device is operating in triode (or linear) region. W 1 Current in Triode Region: I C V V V V 2 L 2 D n ox GS TH DS DS Terminology: Aspect Ratio W L Overdrive Voltage Effective Voltage V GS V TH V eff 23

Long Channel Current Equations - 5 For very small V DS (deep Triode Region): I D can be approximated to be a linear function of V DS. The device resistance will be independent of V DS and will only depend on V eff. The device will behave like a variable resistor If V DS 2V GS V TH : W I D n C ox V GS V TH L V DS V DS 1 R ON I W D n C ox V GS V TH L 24

Long Channel Current Equations - 6 Increasing V DS causes the channel to acquire a tapered shape. Eventually, as V DS reaches V GS V TH the channel is pinched off at the drain. Increasing V DS above V GS V TH has little effect (ideally, no effect) on the channel s shape. Microelectronic Circuits, 2004 Oxford University Press When V DS is more than V GS V TH the channel is pinched off, and the horizontal electric field produces a current. 25

Long Channel Current Equations - 7 If V DS > V GS V TH, the transistor is in saturation (active) region, and the channel is pinched off. L' I D dx x0 V GS V TH WC ox n [V GS V ( x) V TH ]dv V 0 1 W I C (V V 2 L' ) 2 D n ox GS TH Let s, for now, assume that L =L. The fact that L is not equal to L is a second-order effect known as channel-length modulation. Since I D only depends on V GS, MOS transistors in saturation can be used as current sources. 26

Long Channel Current Equations - 8 Current Equation for NMOS: I D I DS 0 n 1 2 ; if V GS V TH (Cut off ) W C ox V GS V TH V DS ; if V GS V TH, V DS 2(V GS V TH ) (Deep Triode) L V 1 V V V 2 n ox GS TH DS DS GS TH DS GS TH C W ; if V V, V V V (Triode) L 2 W ) 2 n Cox (VGS V TH ; if V GS V TH, VDS V GS V TH (Saturation ) L 27

Long Channel Current Equations - 9 Current Equation for PMOS: I D I SD 0 ; if V SG V TH (Cut off ) W p C ox V SG V TH V SD ; if V SG V TH, V SD 2(V SG V TH ) (Deep Triode) L W 1 V V 2 p Cox V SG V TH SD SD ; if V SG V TH, V SD VSG V TH (Triode) L 2 1 W ) 2 p Cox (V SG V TH ; if V SG V TH, V SD VSG V TH (Saturation ) 2 L 28

Regions of Operation - 1 Regions of Operation: Cut-off, triode (linear), and saturation (active or pinch-off) Microelectronic Circuits, 2004 Oxford University Press Once the channel is pinched off, the current through the channel is almost constant. As a result, the I-V curves have a very small slope in the pinch-off (saturation) region, indicating the large channel resistance. 29

Regions of Operation - 2 The following illustrates the transition from pinch-off to triode region for NMOS and PMOS devices. For NMOS devices: If V D increases (V G Const.), the device will go from Triode to Pinch-off. If V G increases (V D Const.), the device will go from Pinch-off to Triode. ** In NMOS, as V DG increases the device will go from Triode to Pinch-off. For PMOS devices: If V D decreases (V G Const.), the device will go from Triode to Pinch-off. If V G decreases (V D Const.), the device will go from Pinch-off to Triode. ** In PMOS, as V GD increases the device will go from Pinch-off to Triode. 30

NMOS Regions of Operation: Regions of Operation - 3 Microelectronic Circuits, 2004 Oxford University Press Relative levels of the terminal voltages of the enhancement-type transistor for different regions of operation. NMOS 31

PMOS Regions of Operation: Regions of Operation - 4 Microelectronic Circuits, 2004 Oxford University Press The relative levels of the terminal voltages of the enhancement-type PMOS transistor for different regions of operation. 32

Regions of Operation - 5 Example: For the following circuit assume that V TH =0.7V. When is the device on? What is the region of operation if the device is on? Sketch the on-resistance of transistor M 1 as a function of V G. 33

Transconductance - 1 The drain current of the MOSFET in saturation region is ideally a function of gate-overdrive voltage (effective voltage). In reality, it is also a function of V DS. It makes sense to define a figure of merit device converts the voltage to current. that indicates how well the Which current are we talking about? What voltage is in the designer s control? What is this figure of merit? g m I D V GS V DS Const. 34

Transconductance - 2 Example: Plot the transconductance of the following (assume V b is a constant voltage). circuit as a function of V DS Transconductance in triode: W V GS 1 2 g m n C ox V TH V DS VGS L 2 W n C ox V DS L Transconductance in saturation: V DS 1 C W g ) 2 m n ox (V GS VTH V GS 2 L V DS Const. W n C ox (V GS V TH ) L V DS Const. Moral: Transconductance drops if the device enters the triode region. 35

Transconductance - 3 Transconductance, g m, in saturation: 2 I g C W (V V ) 2 C W I D m n ox GS TH n ox D L L V GS VTH If the aspect ratio is constant: g m depends linearly on (V GS - V TH ). Also, g m depends on square root of I D. If I D is constant: g m is inversely proportional to (V GS - V TH ). Also, g m depends on square root of the aspect ratio. If the overdrive voltage is constant: g m depends Also, g m depends linearly on the aspect ratio. linearly on I D. 36

Second-Order Effects (Body Effect) Substrate Voltage: So far, we assumed that the bulk and source of the transistor are at the same voltage (V B =V S ). If V B >V s, then the bulk-source PN junction will be forward the device will not operate properly. If V B <V s, the bulk-source PN junction will be reverse biased. the depletion region widens, and Q dep increases. V TH will be increased (Body effect or Backgate effect). biased, and It can be shown that (what is the unit for : 2 q si N sub VTH VTH 0 2 F VSB 2 F where C ox 37

Body Effect - 2 Example: Consider the circuit below (assume the transistor is in the active region): If body-effect is ignored, V TH will be constant, and I 1 will only depend on V GS1 =V in -V out. Since I 1 is constant, V in -V out remains constant. V in V out V TH C Const. V in V out V TH C D Conts. In general, I 1 depends on V GS1 - V TH =V in -V out -V TH (and with body effect V TH is not constant). Since I 1 is constant, V in -V out -V TH remains constant: V in V out V TH C Const. V in V out V TH C As V out increases, V SB1 increases, and as a result V TH increases. Therefore, V in -V out Increases. No Body Effect With Body Effect 38

Body Effect - 3 Example: For the following Circuit sketch the drain current of transistor M 1 when V X varies from - to 0. Assume V TH0 =0.6V, =0.4V 1/2, and 2 F =0.7V. 39

Channel Length Modulation - 1 When a transistor is in the saturation region (V DS > the channel is pinched off. V GS V TH ), L 1 W I (VGS V TH ) 2 D n C ox where L' L-L 2 L' 1 1 1 1 1 1 L L' L L L 1 L L L L 1 1 1 L V 1 1 V L DS L L DS L' L L 1 W 2 1 W 2 I D n C ox (V GS V TH ) 2 L' 2 n C ox V GS V TH 1 V DS L The drain current is Assuming we get: The drain current is As I D actually depends on both V GS and V DS, MOS transistors are not ideal current sources (why?). 40

Channel Length Modulation - 2 represents the relative variation in effective length of the increment in V DS channel for a given For longer channels is smaller, i.e., L Transconductance: g m I D V GS V DS Const. In Triode: g C m n ox W L V DS In Saturation (ignoring channel length modulation): W 2 I g C (V V ) 2 C W I D m n ox GS TH n ox D L L V V In saturation with channel length modulation: W 2 I g V ) 1 V 2 C W D m n Cox (V GS TH DS n ox I D 1 VDS L L V V The dependence of I D on V DS is much weaker than its dependence on V GS GS TH GS TH 41

Channel Length Modulation - 3 Example: Given all other parameters constant, plot I D -V DS characteristic for L=L 1 and L=2L 1 In Triode Region: 2 W V GS V THV DS 1 V DS I D n C ox L 2 I Therefore : D W V DS L of an NMOS In Saturation Region: I D 2 L I D 1 n C ox W V GS V TH 2 1 V DS So we get : 1 n C W ox V GS VTH 2 V DS 2 L I Therefore : D W W L L 2 V DS Changing the length of the device from L 1 to 2L 1 will flatten the I D -V DS curves (slope will be divided by two in triode and by four in saturation). Increasing L will make a transistor a better current source, while degrading its current capability. Increasing W will improve the current capability. 42

Sub-threshold Conduction If V GS < V TH, the drain current is not zero. The MOS transistors behave similar to BJTs. V BE I I e In BJT: C S V T I I In MOS: D 0 V GS e V T As shown in the figure, in MOS transistors, the drain current drops by one decade for approximately each 80mV of drop in V GS. In BJT devices the current drops faster (one decade for approximately each 60mv of drop in V GS ). This current is known as sub-threshold or weak-inversion conduction. 43

CMOS Processing Technology Top and side views of a typical CMOS process 44

CMOS Processing Technology Different layers comprising CMOS transistors 45

Photolithography (Lithography) Used to transfer circuit layout information to the wafer 46

Typical Fabrication Sequence 47

Self-Aligned Process Why source and drain junctions are formed after the gate oxide and polysilicon layers are deposited? 48

Oxide spacers and silicide Back-End Processing 49

Back-End Processing Contact and metal layers fabrication 50

Back-End Processing Large contact areas should be avoided to minimize the possibility of spiking 51

MOS Layout - 1 It is beneficial to have some insight into the layout of the MOS devices. When laying out a design, there are many important parameters we need to pay attention to such as: drain and source areas, interconnects, and their connections to the silicon through contact windows. Design rules determine the criteria that a circuit layout must meet for a given technology. Things like, minimum length of transistors, minimum area of contact windows, 52

MOS Layout - 2 Example: Figures below show a circuit with a suggested layout. The same circuit can be laid out in different ways, producing different electrical parameters (such as different terminal capacitances). 53

Device Capacitances - 1 The quadratic model determines the DC behavior of a MOS transistor. The capacitances associated with the devices are important when studying the AC behavior of a device. There is a capacitance between any two terminals of a MOS transistor. So there are 6 Capacitances in total. The Capacitance between Drain and Source is negligible (C DS =0). These capacitances will depend on the region of operation (Bias values). 54

The following will be used terminals: 1. 2. Oxide Capacitance: Device Capacitances - 2 Depletion Capacitance: to calculate the capacitances between C 1 W L C ox C C 2 dep, W L ox Cox tox q si N su b 4 F 3. Overlap Capacitance: C 3 C 4 C ov W L D C ox C fringe 4. Junction Capacitance: Sidewall Capacitance: C jsw Bottom-plate Capacitance: C C 5 C 6 C j C jsw j C ju n C j0 V 1 R B m 55

In Cut-off: 1. 2. 3. Device Capacitances - 3 C GS : is equal to the overlap capacitance. C GD : is equal to the overlap capacitance. C GS C GD C ov C 3 C ov C 4 C channel-bulk C GB : is equal to C gate-channel = C 1 in series with = C 2. 4. C SB : is equal to the junction capacitance between source and 5. bulk. C DB : is equal to the junction capacitance between source and bulk. C SB C 5 C DB C 6 56

Device Capacitances - 4 In Triode: The channel isolates the gate from the substrate. This means that if V G changes, the charge of the inversion layer are supplied by the drain and source as long as V DS is close to zero. So, C 1 is divided between gate and drain terminals, and gate and source terminals, and C 2 is divided between bulk and drain terminals, and bulk and source terminals. 1. 2. 3. 4. 5. C GS : C GD : C C GS GD C C ov ov C 1 2 C 1 2 C GB : the channel isolates the gate from the substrate. C GB 0 C SB : C DB : C C C SB 5 C DB 6 C 2 2 C 2 2 57

In Saturation: Device Capacitances - 5 The channel isolates the gate from the substrate. The voltage across the channel varies which can be accounted for by adding two equivalent capacitances to the source. One is between source and gate, and is equal to two thirds of C 1. The other is between source and bulk, and is equal to two thirds of C 2. 1. C C C 2 C GS : GS ov 1 3 2. C GD : C GD C ov 3. C GB : the channel isolates the gate from the substrate. C GB 0 4. C SB : C C 2 SB 5 C2 3 5. C DB : C DB C 6 58

Device Capacitances - 6 In summary: Cut-off Triode Saturation C GS Cov C C 1 C 2 C ov 2 3 ov 1 C GD C ov C ov C 2 1 C ov C GB C 1 C 2 C GB C 1 0 0 C 1 C 2 C SB C C C 2 2 C 2 C 3 5 5 5 2 C DB C 6 C C 2 6 2 C 6 SM 59

Importance of Layout Example (Folded Structure): Calculate the gate resistance of the circuits shown below. Folded structure: Decreases the drain capacitance Decreases the gate resistance Keeps the aspect ratio the same 60

Passive Devices Resistors 61

Passive Devices Capacitors: 62

Passive Devices Capacitors 63

Passive Devices Inductors 64

Latch-Up Due to parasitic bipolar transistors in a CMOS process 65

Small Signal Models - 1 Small signal model is an approximation of the large-signal model around the operation point. In analog circuits most MOS transistors are biased in saturation region. In general, I D is a function of V GS, V DS, and V BS. We can use this Taylor series approximation: TaylorExpansion : I I I D V I D V I D V V GS V DS V BS D D0 GS DS BS second order terms I D I V D I V D V I V g V DS g V V GS V DS V BS r o D GS DS BS m GS mb BS 66

Current in Saturation: Small Signal Models - 2 1 W 2 1 W I D n C ox (V GS V TH ) n C ox 2 L' 2 L I I D D V I D I V V BS V V V Taylor approximation: D GS DS Partial Derivatives: I D V GS GS DS BS W n C ox (V GS V TH ) 1 V DS g L m 2 V GS V TH 1 V DS I D 1 W 1 C ) 2 n ox (VGS VTH I D V DS 2 L r o I D I D V TH W C ox (V GS V ) 1 V DS n TH V BS V TH V BS L 2 2 F V SB g m g m g mb 2 2 F V SB 67

Small-Signal Model: Small Signal Models - 3 id g m v GS v DS r o g mb v BS Terms, g m v GS and g mb v BS, can be modeled by dependent sources. These terms have the same polarity: increasing v G, has the same effect as increasing v B. The term, v DS /r o can be modeled using a resistor as shown below. 68

Small Signal Models - 4 Complete Small-Signal Model with Capacitances: Small signal model including all the capacitance (qualitative) analysis of even a few-transistor circuit makes the intuitive difficult! Typically, CAD tools are used for accurate circuit analysis For intuitive analysis we try to find a simplest model that can the role of each transistor with reasonable accuracy. represent 69

Circuit Impedance - 1 It is often useful to determine the impedance of specific pair of terminals. a circuit seen from a The following is the recipe to do so: 1. 2. 3. Connect a voltage source, V X, to the port. Suppress all independent sources. Measure or calculate I X. R X V X I X 70

Example: Circuit Impedance - 2 Find the small-signal impedance of the following current sources. We draw the small-signal model, which is the same for both circuits, and connect a voltage source as shown below: v X i r g v X m GS o v r X o R X v X i X r o 71

Example: Circuit Impedance - 3 Find the small-signal impedance of the following circuits. We draw the small-signal model, which is the same for both circuits, and connect a voltage source as shown below: v X i g v g v g v g v X m GS mb BS m X mb X r r o v X v X 1 1 1 R r X o i 1 g g g g m mb r X m mb o o 72

Example: Circuit Impedance - 4 Find the small-signal impedance of the following circuit. This circuit is known as the diode-connected load, and is used frequently in analog circuits. We draw source as the small-signal model and shown below: v X v X i g v g v v 1 g X m GS m X X m r r r o o o v X 1 1 R r X o i 1 g X m g m r o connect the voltage If channel length modulation is ignored (r o = ) we get: 1 1 1 R r X o g g g m m m 73

Circuit Impedance - 5 Example: Find the small-signal impedance of the following circuit is a diode-connected load with body effect. circuit. This v X i g v g v g v g v X m GS mb BS m X mb X r r o v 1 g g X m mb ro v X 1 1 1 1 R r r X o o i 1 g g g g g g m mb r X m mb m mb o v X o If channel length modulation is ignored (r o = ) we get: 1 1 1 1 1 R r X o g g g g g g g g m mb m mb m mb m mb 74

Equivalent Transconductance - 1 Recall that the transconductance of a transistor was a a figure of merit that indicates how well the device converts a voltage to current. g m I D V V Const. GS DS It is sometimes useful to define the equivalent transconductance of a circuit as follows: I G OUT m V V Const. IN OUT The following is a small-signal block diagram of an arbitrary circuit with a Norton equivalent at the output port. We notice that: V OUT =Constant so v OUT =0 in the small signal model. G m i OUT v IN v 0 OUT 75

Example: Equivalent Transconductance - 2 Find the equivalent transconductance of an NMOS transistor in saturation from its small-signal model. i g v g v OUT m GS m IN G m i OUT v IN g m 76

Example: Equivalent Transconductance - 3 Find the equivalent transconductance of the following circuit when the NMOS transistor in saturation. v v v v i R IN GS S GS OUT S v S i R OUT i g v g v g (v i R ) g i R OUT m GS mb BS m IN OUT S mb OUT S r r O R S i 1 g R g R g v OUT m S mb S m IN ro i g g r OUT m m O G m v R 1 g R g R S r r g R g R R IN O O m S mb S S m S mb S r O O S 77

Short-Channel Effects Threshold Reduction Drain-induced barrier lowering (DIBL) Mobility degradation Velocity saturation Hot carrier effects Substrate current Gate current Output impedance variation 78

Threshold Voltage Variation in Short Channel Devices The Threshold of transistors fabricated on the same chip decreases as the channel length decreases. Intuitively, the extent of depletion regions associated with drain and source in the channel area, reduces the immobile charge that must be imaged by the charge on the gate. 79

Drain-Induced Barrier Lowering (DIBL) When the channel is voltage increases the short, the drain channel surface potential, lowering the barrier to flow charge from source (think of increased electric field) and therefore, decreasing the threshold. 80

Effects of Velocity Saturation Due to drop in mobility at high electric fields (a) Premature drain current saturation and (b) reduction in g m 81

Hot Carrier Effects Short channel devices may experience high lateral drain-source electric field Some carriers that make it to drain have high velocity (called hot carriers) Hot carriers may hit silicon atoms at high speed and cause impact ionization The resulting electron and holes are absorbed by the drain and substrate causing extra drain-substrate current Really hot carriers may be injected into gate oxide and flow out of gate causing gate current! 82

Output Impedance Variation Recall the definition of. 83

Output Impedance Variation in Short-Channel Devices 84