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1 Lecture-2 Date: NMOS I/V Characteristics Discussion on I/V Characteristics MOSFET Second Order Effect

2 NMOS I-V Characteristics ECE315 / ECE515 Gradual Channel Approximation: Cut-off Linear/Triode Pinch-off/Saturation Assumptions: V SB = 0 ; V T is constant along the channel ; E x dominates E y need to consider current flow only in the x direction Cutoff Mode: 0 V GS V T ; I DS(cutoff) =0 This relationship is simple if MOSFET is in cutoff, drain current is simply zero! Linear Mode: V GS V T, 0 V DS V D(SAT) V DS V GS V T The channel reaches the drain. V c (x): Channel voltage with respect to the source at position x. Boundary Conditions: V c x = 0 = V S = 0; V c x = L = V DS

3 Linear Mode (Contd.) Q d : the charge density along the direction of current = WC ox [V GS V T ] where, W = width of the channel and WC ox is the capacitance per unit length The channel potential varies from 0 at source to V DS at the drain: Q d x = WC ox V GS V c (x) V T, where, V c (x) = channel potential at x. Subsequently we can write: I D x = Q d x. v, where, v = velocity of charge (m/s) v = μ n E ; where, μ n = mobility of charge carriers (electron) dv E = electric field in the channel given by: Ex ( ) dx dv Therefore, I D( x) WCox VGS Vc ( x) VT n dx

4 Linear Mode (Contd.) xl VV DS ECE315 / ECE515 Applying the boundary conditions for V c (x) we can write: I ( x) I I. dx WC V V ( x) V. dv D D D ox GS T n x0 V0 Simplification gives the drain current in linear mode as: 2 I D ncox 2VGS VT VDS VDS 2 L Then, I C V V 2 L 2 D,max n ox GS T The I D,max occurs at, V DS = V GS -V T, [how/why?] called overdrive voltage Home Assignment # 0 Observations: I D is dependent on constant of technology (μ n C ox ), the device dimensions (W and L), and the gate and drain potentials with respect to the source W For V DS << 2(V GS -V T ): I D ncox VGS VT VDS Linear function of V L DS

5 Linear Mode (Contd.) For small values of V DS, the drain current can be thought of as a straight line implying that the path from source to drain can be represented by a linear resistor support of earlier assumption R DS V I DS D 1 C W V L V n ox GS T MOSFET transistor operate as a resistor whose value is controlled by overdrive voltage

6 Pinch-off point (Edge of Saturation): V GS V T, V DS = V D(SAT) The channel just reaches the drain but with zero inversion at the drain Electrons start to drift from the channel to the drain The drain current is given by: I I C V V 2 L Saturation Mode: V GS V T, V DS V GS V T After pinch-off, the I D saturates i.e, is relatively independent of V DS : I C V V 2 L 2 D, sat n ox GS T 2 D D,max n ox GS T 2 I D ncox 2VGS VT VDS VDS 2 L On the right of this parabolic curve I D, sat ncox VGS VT 2 L 2

7 Saturation Mode (Contd.) MOSFET can be used as current source connected between the drain and the source, controlled by V GS. MOSFET in saturation mode produces a current regulated by V GS imperative to define a figure of merit (FOM) that identifies the effectiveness with which the MOSFET can convert voltages in currents the FOM in this scenario is called transcodunctance (g m ). Defined as the change in the drain current divided by the change in the gate-source voltage. I W g C V V D m n ox GS T VGS L V, const. In essence, g m represents the sensitivity of the device. For a high g m, a small change in V GS results in a large change in I D. DS W Other formulations of g m : g 2. C I L m n ox D g m V 2I D GS V T

8 Transconductance (g m ) ECE315 / ECE515 Behavior of g m as a function of one parameter while other parameters remain fixed. Home Assignment # 0 Can we define g m in the Channel Resistance for Small V triode/linear region? DS Voltage V DS will be directly proportional to I D, provided that: 1. A conducting channel has been induced. 2. The value of V DS is small. Note for this situation, the MOSFET will be in triode region.

9 Channel Resistance for Small V DS As we increase the value of V DS, the conducting channel will begin to pinch off the current will no longer be directly proportional to V DS. Specifically, there are two phenomena at work as we increase V DS while in the triode region: 1. Increasing V DS will increase the potential difference across the conducting channel leads to proportional increase in I D. 2. Increasing V DS will decrease the conductivity of the induced channel leads to decrease in I D. There are two physical phenomena at work as we increase V DS, and there are two terms in the triode drain current equation! 2 I D ncox 2VGS VT VDS VDS 2 L W I C V V V C V L 2 L 2 D n ox GS T DS n ox DS Where: W I C V V V L D1 n ox GS T DS I I I D D1 D2 I C V 2 L 2 D2 n ox DS

10 W I C V V V L D1 n ox GS T DS ECE315 / ECE515 It is evident that this term describes the first phenomenon: 1. Increasing V DS will increase the potential difference across the conducting channel leads to proportional increase in I D. I D1 V GS V T In other words, this first term would accurately describe the relationship between I D and V DS if the MOSFET induced channel behaved like a resistor! it means the second term doesn t allow it to behave like a perfect resistor. V DS I C V 2 L 2 D2 n ox DS It is apparent that I D2 is proportional to V DS squared!! Moreover, the minus sign means that as V DS increases, I D2 will actually decrease! This behavior is nothing like a resistor what the heck is going on here?? I D2 V GS V T V DS

11 This second term essentially describes the result of the second phenomena: 2. Increasing V DS will decrease the conductivity of the induced channel leads to decrease in I D. Now let s add the two terms I D1 and I D2 together to get the total triode drain current I D : It is apparent that the second term I D2 works to I D reduce the total drain current from its resistorlike value I D1. This of course is physically due I D1 to the reduction in channel conductivity as V DS increases. I D V GS V T V DS Q: But look! It appears to me that for small values of V DS, the term I D2 is very small, and thus I D I D1 (when V DS is small)! I D2 A: Absolutely true! Recall this is consistent with our earlier discussion about the induced channel the channel conductivity begins to significantly degrade only when V DS becomes sufficiently large!

12 Channel Resistance (contd.) Thus, we can conclude: I D I D1 ncox VGS VT VDS For small V DS Moreover, we can (for small V DS ) approximate the induced channel as a resistor R DS of value R DS = V DS W L R DS W : ncox VGS VT I D L 1 For small V DS Q: I ve just about had it with this for small V DS nonsense! Just how small is small? How can we know numerically when this approximation is valid? A: Well, we can say that this approximation is valid when I D2 is much smaller than I D1 (i.e., I D2 is insignificant). Mathematically, we can state as: ID2 ID 1 2 W C V C V V V 2 L L V 2 V V n ox DS n ox GS T DS DS GS T

13 Channel Resistance (contd.) ECE315 / ECE515 Thus, we can approximate the induced channel as a resistor R DS when V DS is much less than the twice the excess gate voltage. R DS 1 C W V L V n ox GS T For 2 V V V DS GS T Q: There you go again! The statement V DS 2(V GS V T ) is only slightly more helpful than the statement when V DS is small. Precisely how much smaller than twice the excess gate voltage must V DS be in order for our approximation to be accurate? A: We cannot say precisely how much smaller V DS needs to be in relation to 2(V GS V T ) unless we state precisely how accurate we require our approximation to be! For example, if we want the error associated with the approximation I D I D1 to be less than 10%, we find that we require the voltage V DS to be less than 1/10 the value 2(V GS V T ). In other words, if: 2VGS VT we find then that I VGS V D2 is I D1 T I D2 VDS less than 10% of I D1 :

14 Channel Resistance (contd.) ECE315 / ECE515 This 10% error criteria is a typical rule-of thumb for many approximations in electronics. However, this does not mean that it is the correct criteria for determining the validity of this (or other) approximation. For some applications, we might require better accuracy. For VGS VDS example, if we require less than 5% error, we would find that: 10 It is important to note that we should use these approximations when we can it can make our circuit analysis much easier! V T See, the thing is, you should use these approximations whenever they are valid. They often make your circuit analysis task much simpler

15 Second Order Effect - Channel Length Modulation I have been saying that for a MOSFET in saturation, the drain current is independent of the drain-to-source voltage V DS i.e. I C V V 2 L 2 In reality, this is only approximately true! D n ox GS T Let us look at operation of NMOS in saturation mode: Observations: The pinch-off point moves towards the source with the increase in V DS Channel length reduces Channel resistance decreases This modulation of channel length (L) by V DS is known as channel-length modulation, and leads to slight dependence of I D on V DS. The drain current in saturation mode is: I C V V 2 L actually 1 2 W L D, sat n ox GS T varies with V DS The decrease in channel length with increase in V DS essentially increases the drain current I D

16 Second Order Effect - Channel Length Modulation (contd.) If ΔL = L-L 1 then: (1 VDS ) L L 1 L L L 1 L 1 V DS L L λ: channel length modulation coefficient (usually less than 0.1) Therefore the drain current in saturation mode becomes: I 2 D, sat ncox VGS VT (1 VDS ) 2 L λ is a MOSFET device parameter with units of 1/V (i.e., V -1 ). Typically, this value is small (thus the dependence on V DS is slight), ranging from to 0.02 V -1.

17 Second Order Effect - Channel Length Modulation (contd.) Often, the channel-length modulation parameter λ is expressed as the Early Voltage V A, which is simply the inverse value of λ. The parameter V A is set at the time of fabrication and hence the circuit designers can t alter it at circuit/system design stage. V A = 1 λ The drain current for a MOSFET in saturation can likewise be expressed as: I D ncox VGS VT (1 VDS ) 2 L V 1 2 L V 2 I C V V 2 D n ox GS T Now, let s define a value I DI, which is simply the drain current in saturation if no channel-length modulation actually occurred in other words, the ideal value of the drain current: DS A I C V V 2 L 2 DI n ox GS T

18 Second Order Effect - Channel Length Modulation (contd.) Thus, we can alternatively write the drain current in saturation as: I D I DI V V DS 1 This explicitly shows how the drain current behaves as a function of voltage V DS. A I D I DI V V 1 DS A We can interpret the value V DS VA as the percent increase in drain current I D over its ideal (i.e., no channel length modulation) saturation value Now, let s introduce a third way (i.e. in addition to, λ and V A ) to describe the extra current created by channel-length modulation. Define the Drain Output Resistance r o : r o V I I 1 A Using this definition, we can write the saturation drain current expression as: I D I DI V V DS 1 A V I D ncox VGS VT 2 L r DS o DI DI

19 Second Order Effect - Channel Length Modulation (contd.) Thus, we interpret the extra drain current (due to channel length modulation) as the current flowing through a drain output resistor r o. Finally, there are three important things to remember about channel-length modulation: The values λ and V A are MOSFET device parameters, but drain output resistance r o is not (r o is dependent on I DI ). Often, we neglect the effect of channel-length modulation, meaning that we use the ideal case for saturation: I D = I DI = μ n C ox (V GS V T ) 2. Effectively, we assume that λ = 0, meaning that V A = and r o = (i.e., not V A = 0 and r o = 0). The drain output resistance r o is not the same as channel resistance R DS. The two are different in many, many ways.

20 Second Order Effect - Channel Length Modulation (contd.) For a longer channel length, the relative change in L and therefore in I D for a given change in V DS is smaller. To minimize channel length modulation, smaller length transistors should be avoided. Q: Any idea about limitation of long channel devices? Home Assignment # 0

21 Second Order Effect Body Effect In discrete circuit usually there is no body effect as the body is connected to the source terminal. In integrated circuit, there are thousands or millions of MOSFET source terminals and there is only one Body (B) the silicon Substrate. Thus, if we were to tie (connect) all the MOSFET source terminals to the single body terminal, we would be connecting all the MOSFET source terminals to each other! This would almost certainly result in a useless circuit! Therefore, for integrated circuits, the MOSFET source terminals are not connected to the substrate body. Actually, the substrate is connected to the most negative power supply for NMOS circuit for achieving the desired functionality from the device. In such a scenario, what happens if the bulk voltage drops below the source voltage? Now the voltage V SB (voltage source-to-body) is not necessarily equal to zero (i.e., V SB 0 ). Thus, we are back to a four-terminal MOSFET device. There are many ramifications of this body effect; perhaps the most significant is with regard to the threshold voltage V T.

22 Second Order Effect Body Effect ECE315 / ECE515 To understand, let us assume V S = V D = V B = 0, and V G is somewhat less than V T. A depletion region forms but no inversion layer exists. As V B becomes negative, more holes get attracted to the substrate which leaves a larger negative charge behind and as a result the depletion region becomes wider. Wider depletion region The wider depletion region leads to increase in threshold voltage given by: V V 0 2 V 2 T T F SB F where γ and Φ F are MOSFET device parameters and are essentially process dependent. Note the value V T0 is the value of the threshold voltage when V SB = 0 the value V T0 is simply the value of the device parameter V T that we have been calling the threshold voltage up till now, i.e., without body effect.

23 Threshold Voltage (V) Second Order Effect Body Effect ECE315 / ECE515 It is thus evident that the term γ 2Φ F + V SB 2Φ F simply expresses an extra value added to the ideal threshold voltage V T0 when V SB 0. Questions Can a circuit designer control threshold voltage? Body effect is desirable or undesirable? Home Assignment # 0 Substrate Doping (cm 3 )

24 Second Order Effect Body Effect (contd.) Effect of body effect on Transconductance (g m ). Let us check the sensitivity of I DS to V SB g m Simplification Yields: V T V T F VBS VSB 2 W V gmb ncox VGS VT 0 L V 2 V 1 2 SB g g g 2 2 V T BS mb m m F SB η=1/3 to 1/4, bias dependent Thus, body effect has potential to alter (reduce) transconductance and therefore can impact device performance adversely For many cases, we find that this Body Effect is relatively insignificant, so we will (unless otherwise stated) ignore the Body Effect. However, do not conclude that the Body Effect is always insignificant it can in some cases have a tremendous impact on MOSFET circuit performance!

25 Second Order Effect Subthreshold Conduction For V GS V T, a week inversion layer still exists and some current flows from D to S. Even for V GS <V T, I D is finite subthreshold conduction For V DS greater than roughly 200 mv: Nonideality Factor I 0 exp GS V TM = kt/q When V GS is brought to zero, there will be some drain current in the channel. This is clearly unwanted situation as this small current will cause significant power consumption in large circuits D I V V TM

26 Second Order Effect Voltage Limitations Various breakdown occurs if their terminal voltage differences exceed certain limits At high V GS, the gate oxide breaks down irreversibly In short-channel devices, an excessively large V DS can widen the depletion region around the drain so much that it touches that around the source, creating a very large drain current punch through Question Can channel length modulation affect NMOS/PMOS performance in linear/triode operation mode?

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