Lecture 6, ATIK. Switched-capacitor circuits 2 S/H, Some nonideal effects Continuous-time filters

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1 Lecture 6, ATIK Switched-capacitor circuits 2 S/H, Some nonideal effects Continuous-time filters

2 What did we do last time? Switched capacitor circuits The basics Charge-redistribution analysis Nonidealties SC parasitics 155 of 529

3 What will we do today? Switched capacitor circuits with non-ideal effects in mind What should we look out for? What is the impact on system performance, like filters. Continuous-time filters Active-RC Transconductance-C Second-order links Leapfrog filters Mainly overview 156 of 529

4 A list of non-ideal effects in SC System Parasitics Noise OP Offset error Gain Bandwidth (output impedance) Slew rate Noise Switches On-resistance Clock feed through, Charge injection Jitter 157 of 529

5 An example SC accumulator to work on Phase 1: q1 nt =C 1 V 1 nt, q2 nt =C 2 V 2 nt Phase 2: q1 nt =0, q2 nt =C 2 V 2 nt v2 t v 1 t Charge preservation: q2 nt =q2 nt V 2 nt =V 2 nt q1 nt q 2 nt = q1 nt q2 nt = q 2 nt T Laplace transform V 2 ( z) / C 1 V 1 ( z)= C 2 ( 1 z ) V 2 ( z) = 1 V 1 ( z) 1 z of 529

6 Impact of offset error Due to mismatch in differential pair, we will have an offset at the input of our amplifier I = V 2eff I = V 2eff The offset error is in the order of v2 t v 1 t 2 V I V eff eff V eff = = = gm 2ID 2 V eff With the help of gain, we can propagate any mismatch back to the input which results in a constant voltage, v x, on (one of) the input(s) 159 of 529

7 Impact of offset error, cont'd Phase 1 q 1 nt =C 1 v 1 nt v x, q 2 nt =C 2 v 2 nt v x Phase 2 q 1 nt =0, q 2 nt =C 2 v 2 nt v x Charge preservation q1 nt q 2 nt = q1 nt q2 nt v2 t v 1 t q 2 nt =q 2 nt T v 2 nt =v 2 nt T C 1 v 1 n T C 1 v x C 2 v 2 nt C 2 v x =C 2 v 2 nt T C 2 v x vx C 1 /C 2 v x C 1 /C 2 C 1 V 1 z C 2 V 2 z =C 2 V 2 z z V 2 z = V 1 z z 1 z 1 z vx ooops! Output is ramped due to offset! 160 of 529

8 Impact of gain errors Phase 1 q 1 nt =C 1 v 1 nt v 2 nt v 2 nt q nt =C v nt, A0 A0 Phase 2 v nt q 1 nt =0, q 2 nt =C 2 v 2 nt 2 A0 v2 t v 1 t Charge preservation q1 nt q 2 nt = q1 nt q2 nt, q 2 nt =q 2 nt T v 2 nt =v 2 nt T 1 1 C 1 v 1 n T = v 2 nt C 2 1 C 2 1 v 2 nt T A0 A0 A0 161 of 529

9 Impact of gain errors, cont'd Compiled V 2 z = V 1 z C 1 /C 2 1 1/ A0 C 1 /C 2 z 1 A0 1 1 v2 t v 1 t Introduces a gain error and a pole shift (!) Lossy integrator, i.e., a low-pass filter with a DC gain of 1 V 2 (1) 1+ 1/ A0 = = A0 V 1 (1) 1 A of 529

10 Impact of bandwidth The speed (bandwidth) of the OP is given by the unity-gain frequency and feedback factor 1/ 1/ H s = = 1 1 s / p A s A0 v2 t v 1 t 1 / 1/ s s 1 1 A0 p1 ug This means that the output will follow a step response according to: v 2 nt t =v 2 nt V 2 1 e t ug Notice that the feedback factor varies with different phases! 163 of 529

11 Impact of bandwidth, cont'd Phase 2: discharging C 1 is instantaneous. V 2 cannot change either, since charge on C 2 is maintained due to the infinitely fast switch. Phase 1: we re-charge C 1 and settling of V 2 v2 t v 1 t will determine how fast we can do that. A first-order, lazy approximation: v 2 nt =v 2 nt v 1 nt (ideal) v 2 nt t =v 2 n T v 2 nt v 1 nt v 2 nt 1 e t v 2 nt ug (actual) v 2 nt =v 2 nt v 1 nt 1 e ug B 164 of 529

12 Impact of bandwidth, cont'd Compiled (notice the approximation, in reality an additional time-shifted gain component too!): V 2 z C 1 /C 2 B V 1 z z 1 v2 t v 1 t Ideal Less of a problem in this case, in a first-order analysis, it results in a gain error. The ideal-switch assumption and charge preservation forces the accumulation to not be lossy. The clock frequency, f s, is hidden in the equation and if ug 2 f s, the B is a rather small value! 165 of 529

13 Impact of slew rate Slew rate is a non-linear function. The output will now follow: I max v 2 nt t =v 2 nt t CL Within a phase, we are able to reach v 2 t v 1 t I max v 2, max = CL Generic analysis hard, since only large voltage steps are affected If SR is not avoided we have a highly distorted signal! 166 of 529

14 Impact of noise Noise due to a switch in Nyquist band: 2 v C ( f )=4 k T R equiv = 4k T f s C 1 v2 t Noise from operational amplifier: v 2 t v 1 t Given by input-referred noise voltage: 2 v op (f) Noise is sampled, i.e., aliased At the sampling instant, the noise voltage at the input of the OP is sampled. C.f. offset error C 1 /C 2 + V X ( z ) 1 z 167 of 529

15 Impact of noise, cont'd The super function C 1 /C 2 2 S out z = H z S i n z S i n z 1 z 2 The transfer function (in this case) modifies the noise to the output and noise is integrated too. v2 t v 2 t v 1 t Beware! Continuous-time noise vs sampled noise. (When do you measure the noise)? 168 of 529

16 Impact of on-resistance v2 t Similar to limited bandwidth, but now a bit more accurate step-by-step approach: R q1 =C 1 v 1 1 e =C 1 v 1 B B v 2 t v 1 t q 1 R C q1 T =C 1 e =q 1 A A 1 q1 (T ) q 1 (T ) q1 (T + τ)=c 1 + C 1 v 1 (T + τ) B ( ) q1 (T + τ)= A q1 ( τ)+ ( C 1 v 1 (T + τ) q1 ( τ) A ) B= A2 q1 ( τ)+ C 1 v 1 (T + τ) B Laplace Q 1 ( z) z =A Q 1 ( z) z C 1 V 1 ( z ) z B Q 1 ( z)= C 1 V 1 ( z) 2 1 A z of 529

17 Impact of on-resistance, cont'd Charge accumulation and preservation must still hold q1 (nt )+ q 2 (nt )=q1 (nt τ)+ q 2 (nt τ) Be careful with which charge is which... A Q 1 ( z) z C 2 V 2 ( z) z 0.5=Q 1 ( z ) z C 2 V 2 ( z) z 0.5 C 1 V 1 ( z) ( A 1 ) =C 2 V 2 ( z) (1 z) A z v 2 t Finally, we get that small time shift (additional pole, close to the origin) V 2 z C 1 /C 2 B = 2 1 V 1 z 1 A z z 1 v2 t v 1 t Ideal 170 of 529

18 Impact of charge feed-through Channel charge injection qch (nt ) W L C ox ( V x V 1 (nt ) ) v2 t Clock feed-through qcft (nt )=C ol ( V y V 1 (nt ) ) v 2 t v 1 t One signal-dependent part and one constant Gain error C 1 ' =C 1 + Δ C Offset accumulation, W L C ox V x + C ol V y The errors can be reduced to nearly zero using e.g. differential signals, switch dummies, and careful sizing. 171 of 529

19 Impact of jitter Sampling instant will be varying nt '=nt T where T might be a stochastic and/or v2 t signal-dependent component v 1 t =sin t. i.e., v 1 nt =sin nt Taylor v 2 t v 1 t v 1 t v 1 nt T v 1 nt v 1 nt t Example v 1 nt T =sin nt cos T cos nt sin T sin nt T cos nt The higher signal frequency the worse! 172 of 529

20 Conclusions SC building blocks Many, many possible error sources Today we've looked at different ways to model and address them and their different impacts Notable error impacts Offset accumulation Overall gain error Shifted pole (lossy integrator) Parasitic pole ( nonlinear integration) Nonlinearity (distortion) Jitter 173 of 529

21 Continuous-time filters Filters and filtering functions are everywhere... Band-select Anti-aliasing Reconstruction filters 174 of 529

22 Continuous-time filters A general transfer function for a linear system is given by 2 Y s a 0 a 1 s a2 s H s = = X s b 0 b 1 s b2 s 2 Rewrite in its original ODE form Y (s) ( b 0+ b1 s+ b2 s2 + )= X ( s) ( a 0+ a1 s+ a 2 s 2+ ) Invert to get integrations rather than derivations, and scale Y s = X s Y s s s s s Create a recursive set of integrations Y s = 0 X s 1 X s 1 Y s 2 X s 2 Y s s s s 175 of 529

23 Continuous-time filters, flow graph Manipulation Replacing with integrators Feedback, etc. 176 of 529

24 Active-RC Summation of different inputs is done with the resistors, i.e., we are summing up the currents in the virtual ground node: V 1 (s) V 2 ( s) V out ( s) s C L =I 1 ( s)+ I 2 ( s)+ I 3 (s)= + + R1 R2 Which combined gives us the integration V out ( s)= V 1 (s ) V 2 ( s) + s C L R1 s C L R2 v3 v2 v1 R3 C R2 R1 v out 177 of 529

25 Example, first-order pole with active-rc Sum the currents in the virtual ground: V i n s V out s V out s s C 2 = R4 R8 such that V out s R8 / R 4 = V i n s s 1 1/C 2 R 4 R4 R8 178 of 529

26 Example: Tow-Thomas, Biquad,... Second-order link cascaded to form overall transfer function R1 R2 R3 R8 R7 R9 V out R4 R5 R6 Vin Output isolated and buffered with OP 179 of 529

27 Single-pole formed with single RC at output or input 180 of 529

28 Ladder networks Use a reference filter A [db] L2 Ri V out 20 V in C3 RL f [khz] Optimum wrt. sensitivity (exercises in lessons) 181 of 529

29 State-space representation Note the voltages and currents through the ladder network X1 E X3 X2 X 5 =V out X4 E =V i n, X 1=V 1, X 2=R I 2, X 3=V 3, X 4= R I 4, 182 of 529

30 Continuous-time filter implementation Gm-C 183 of 529

31 Leapfrog filters See the handouts for state-space realization and implementations K_0NNN_LN_leapfrogFiltersOH1_A.pdf K_0NNN_LN_leapfrogSynthesisExtra1_A.pdf K_0NNN_LN_leapfrogSynthesisExtra2_A.pdf NTIK_0NNN_LN_switcapHandout_B.pdf 184 of 529

32 What did we do today? Switched capacitor circuits with nonideal effects in mind What should we look out for? What is the impact on system performance, like filters. Continuous-time filters The way forward, and the background to generate the filters. OTA-C, Gm-C, and active-rc 185 of 529

33 What will we do next time? Continuous-time filters Wrap-up and some more conclusions Discrete-time filters Simulation of the continuous-time filters 186 of 529

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